High speed signaling system with adaptive transmit pre-emphasis

ABSTRACT

A signaling system having first and second sampling circuits and an output driver circuit. The first sampling circuit samples a first signal generated by the output driver circuit to determine whether the first signal exceeds a first threshold. The second sampling circuit samples the first signal to determine whether the first signal exceeds a second threshold. The drive strength of the output driver circuit is adjusted based, at least in part, on whether the first signal exceeds the first and second thresholds, and the second threshold is adjusted based, at least in part, on whether the first signal exceeds the second threshold.

FIELD OF THE INVENTION

The present invention relates generally to the field of communications,and more particularly to high speed electronic signaling within andbetween integrated circuit devices.

BACKGROUND

Electrical pulses transmitted on a band-limited signaling path dispersein time as they travel from source to destination. In systems in whichdata is transmitted as a sequence of level-encoded electrical pulses,such time-domain dispersion results in a blending of neighboring pulses;an effect known as dispersion-type inter-symbol interference (ISI).Dispersion-type ISI becomes more pronounced at faster signaling rates,ultimately degrading the signal quality to the point at whichdistinctions between originally transmitted signal levels may be lost.

FIG. 1 illustrates a prior-art signaling system having a transmitter101, signal path 102 and receiver 103. The transmitter includes post-and pre-tap output drivers 109 that mitigate dispersion-type ISI bygenerating dispersion-countering, pre-emphasis signals based onpreviously transmitted values (post-tap data) and thenext-to-be-transmitted data value (pre-tap data), stored in shiftregister elements 107 and 104, respectively. The pre-emphasis signalsare wire-summed with a primary output signal, generated by primaryoutput driver 105, that corresponds to the data value being transmitted.

In a low-noise system, the drive strengths of the post- and pre-tapoutput drivers would theoretically be adjusted based on errors betweenreceiver-sampled signal levels and expected signal levels (e.g., asshown by error indication, “e” at 112) until the pre-emphasis signalsgenerated by the transmitter effect a transfer function (W) that is anexact inverse of the transfer function (P) of the signal path 102,thereby yielding a waveform at the input of receiver 103 that isidentical to the primary output signal (i.e., W*P=1). This effect isillustrated in the waveforms of FIG. 2, which illustrates receivedsignal levels with and without pre-emphasis at 116 and 114,respectively. In practical high-speed signaling systems, however, thetransmitter is usually peak power constrained and therefore unable toprovide the level of pre-emphasis needed to restore received signals tooriginally transmitted levels, illustrated as normalized +/−1 signallevels in FIG. 2. Also, as shown at 118, in a level-encoded signalingprotocol (e.g., pulse amplitude modulation (PAM)), the overallattenuation of the received signal is a function of the transmitted datapattern itself, with low frequency components (e.g., sequences ofsame-level transmissions) having a higher amplitude, approaching the+/−1 levels, than high frequency components (e.g., alternating sequencesof different-level transmissions) which are attenuated to +/−a levels.

Together, the transmitter power constraint and the data-dependentattenuation present a number of challenges in the prior-art signalingsystem 100. A fundamental problem is how to generate the error signalused to adjust the drive strengths of the transmitter output driversconsidering that no known data level can be reached for all datapatterns. That is, if the known reference levels +/−1 cannot be reachedin high-frequency data patterns, attempting to converge to such levelstends to produce non-optimal drive-strength settings from the standpoint of link performance.

One prior-art solution for generating error signals that may be used toupdate the drive strengths of the transmitter output drivers is toprovide a variable gain element, G (shown in dashed outline in FIG. 1),at the receive-side of the signaling path 102. In theory, the gainelement may be used to restore the incoming signal to the desiredsignaling level. While some improvement may be realized by such anapproach, as signaling rates progress deeper into the gigahertz range,signals are often attenuated 10 to 20 db and more. Consequently, thegain-bandwidth product required to restore such high data rate signalsto originally transmitted levels is beyond the capability of mostpractical amplifiers.

In view of the challenges involved in dynamically updating drivestrengths of transmit-side output drivers, many system designers opt fora simpler approach, setting the drive strengths based on empiricalresults obtained in particular system configurations. While such staticdrive strength settings work well in many systems, non-optimal settingsoften result in systems which are subject to post-productionconfiguration changes (e.g., adding modules, circuit boards or othercomponents that affect signaling system characteristics), and systemsthat are sensitive to process variations and to changes in environmentalfactors such as voltage and temperature.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention is illustrated by way of example, and not by wayof limitation, in the figures of the accompanying drawings and in whichlike reference numerals refer to similar elements and in which:

FIG. 1 illustrates a prior-art signaling system;

FIG. 2 illustrates an idealized amplification of a channel-attenuatedwaveform;

FIG. 3 illustrates a signaling system according to an embodiment of theinvention;

FIG. 4 illustrates a relationship between clock and data signals in oneembodiment of the signaling system of FIG. 3;

FIG. 5 illustrates an embodiment of a differential output driver thatmay be used to implement each of the output drivers shown in FIG. 3;

FIG. 6 illustrates waveforms that correspond to a substantiallyflattened channel response obtained in the signaling system of FIG. 3;

FIG. 7 illustrates an adaptive module according to an embodiment of theinvention;

FIG. 8 illustrates a power scaling circuit according to an embodiment ofthe invention;

FIG. 9 illustrates a power scaling circuit according to anotherembodiment of the invention;

FIG. 10 illustrates an embodiment of a differential sampler that may beused to implement the data sampler and adaptive sampler shown in FIG. 3;

FIG. 11 illustrates an embodiment of a current DAC that may be used toimplement the current DACs within the sampler of FIG. 10;

FIG. 12 illustrates an alternative embodiment of a sampler that may beused to implement the data sampler and adaptive sampler shown in FIG. 3;

FIG. 13 is a canonical diagram of a channel and receive-side equalizerthat may be used to adaptively determine a set of equalizer tap weights;

FIGS. 14A and 14B are canonical diagrams that illustrate adaptivedetermination of transmit pre-emphasis tap weights using a two-phaseupdate operation;

FIG. 15 is a flow diagram of the two-phase tap weight update operationdescribed in reference to FIGS. 14A and 14B;

FIG. 16 is a canonical diagram that illustrates adaptive determinationof transmit pre-emphasis tap weights using a single-phase updateoperation;

FIG. 17 is a flow diagram of the single-phase tap weight updateoperation described in reference to FIG. 16;

FIG. 18 illustrates a multi-sample receiver according to an embodimentof the invention;

FIG. 19 illustrates a multi-level signaling system according to anembodiment of the invention;

FIG. 20 illustrates an embodiment of a multi-level output driver thatmay be used to implement each of the multi-level output drivers shown inFIG. 19;

FIG. 21 illustrates an exemplary signal encoding protocol used withinthe multi-level signaling system of FIG. 19,

FIG. 22 illustrates an adaptive module according to another embodimentof the invention;

FIG. 23 illustrates an embodiment of a multi-sample, multi-levelreceiver that recovers both data and clocking information from anincoming multi-level signal;

FIG. 24 illustrates possible signal transitions between successive 4-PAMdata transmissions received by the multi-level receiver of FIG. 23.

FIG. 25 illustrates an embodiment of a clock recovery circuit that maybe used to implement the clock recovery circuit shown in FIG. 23;

FIG. 26 illustrates a double-data-rate, multi-sample receiver accordingto an embodiment of the invention;

FIG. 27 illustrates a portion of the receiver of FIG. 26 in greaterdetail;

FIG. 28 illustrates a multi-sample, multi-level receiver according to anembodiment of the invention;

FIG. 29 illustrates an error trap zone and its relationship with anexemplary 2-PAM data waveform;

FIG. 30 illustrates a multi-sample receiver that generates a trapthreshold according to an embodiment of the invention;

FIG. 31 illustrates an error trap zone and its relationship with anexemplary 4-PAM data waveform; and

FIG. 32 illustrates a multi-sample, multi-level receiver that generatesa trap threshold according to an embodiment of the invention.

DETAILED DESCRIPTION

In the following description and in the accompanying drawings, specificterminology and drawing symbols are set forth to provide a thoroughunderstanding of the present invention. In some instances, theterminology and symbols may imply specific details that are not requiredto practice the invention. For example, the interconnection betweencircuit elements or circuit blocks may be shown or described asmulti-conductor or single conductor signal lines. Each of themulti-conductor signal lines may alternatively be single-conductorsignal lines, and each of the single-conductor signal lines mayalternatively be multi-conductor signal lines. Signals and signalingpaths shown or described as being single-ended may also be differential,and vice-versa. Similarly, signals described or depicted as havingactive-high or active-low logic levels may have opposite logic levels inalternative embodiments. As another example, circuits described ordepicted as including metal oxide semiconductor (MOS) transistors mayalternatively be implemented using bipolar technology or any othertechnology in which a signal-controlled current flow may be achieved.With respect to terminology, a signal is said to be “asserted” when thesignal is driven to a low or high logic state (or charged to a highlogic state or discharged to a low logic state) to indicate a particularcondition. Conversely, a signal is said to be “deasserted” to indicatethat the signal is driven (or charged or discharged) to a state otherthan the asserted state (including a high or low logic state, or thefloating state that may occur when the signal driving circuit istransitioned to a high impedance condition, such as an open drain oropen collector condition). A signal driving circuit is said to “output”a signal to a signal receiving circuit when the signal driving circuitasserts (or deasserts, if explicitly stated or indicated by context) thesignal on a signal line coupled between the signal driving and signalreceiving circuits. A signal line is said to be “activated” when asignal is asserted on the signal line, and “deactivated” when the signalis deasserted. Additionally, the prefix symbol “/” attached to signalnames indicates that the signal is an active low signal (i.e., theasserted state is a logic low state). A line over a signal name (e.g.,‘{overscore (<signal name>)}’) is also used to indicate an active lowsignal. The term “terminal” is used to mean a point of electricalconnection. The term “exemplary” is used to express but an example, andnot a preference or requirement.

Signaling systems having a multiple-output driver transmit circuit aredisclosed in various embodiments. In one embodiment, the drive strengthsof output drivers within the transmit circuit are adaptively adjustedconcurrently with adaptive determination of a target, receive-sidesignal level. Thus, even as adaptive determination of the target signallevel is ongoing, the target signal level is compared with receivedsignals to generate error signals that are used, in turn, to adjust thedrive strengths of the output drivers. By this operation, a targetsignal level is determined and used to establish drive strength valuesthat yield a substantially flattened channel response todifferent-frequency transmit data patterns.

In one embodiment, the error signals that result from comparison ofreceived signals with the target signal level are input to a circuitthat generates updated drive strength values, referred to herein as tapweights, in a manner that converges to a least-mean-square (LMS) error.In an alternative embodiment, the target signal level is used toestablish a trap range, with signals falling within the trap range beingused to update the drive strength values. In either embodiment, afterbeing updated, the set of drive strength values may be scaled accordingto the transmit circuit power constraint. By this operation, theadaptive determination of the target signal level converges to a levelthat corresponds to the peak (or average) power available to the signaltransmitter. Thus, a target level that corresponds to a substantiallyflattened frequency response at the peak or average power available tothe signal transmitter is, in effect, learned by the system and used asan error reference for continued adjustment of output driver drivestrengths.

In one implementation, a Taylor series approximation is used to simplifythe power scaling of the drive strength values, enabling the scalingoperation to be carried out in a relatively small logic circuit. Inalternative implementation, drive strength values for pre- and post-tapoutput drivers of the transmit circuit are first updated and the drivestrength of the data driver adjusted up or down to maintain the overalltransmit power level within a predefined range.

In yet other embodiments of the invention, DC offsets within individualsamplers of the receive circuit are adaptively canceled; multiplexingcircuitry is provided to enable one or more samplers within the receivecircuit to be temporarily removed from service and replaced by anothersampler; and single- and two-phase techniques are applied to generatedrive strength update values. These and other features and aspects ofthe invention are disclosed below.

Signaling System Overview

FIG. 3 illustrates a signaling system 200 according to an embodiment ofthe invention. The signaling system 200 includes a multi-output drivertransmitter 201 (referred to herein as a multi-tap transmitter) andmulti-sample receiver 209 coupled to one another via a high-speed signalpath 202. In many of the embodiments described herein, the signal path202 is a differential signal path having a pair of component signallines to conduct differential signals generated by the transmitter 201.In all such embodiments, the signal path 202 may alternatively besinge-ended (i.e., single conductor path) for transmission ofsingle-ended signals generated by the transmitter 201. The signal path202 may be formed in multiple segments disposed on different layers of acircuit board and/or multiple circuit boards. For example, in oneapplication the signal path 202 extends between two backplane-mounteddaughterboards, and includes a printed trace segment on the backplanethat extends between daughterboard connectors and counterpart tracesegments on the daughterboards coupled to one another, via thedaughterboard connectors and the backplane trace segment. Thetransmitter 201 and receiver 209 are implemented in respectiveintegrated circuit (IC) devices that are mounted on a common circuitboard or different circuit boards (e.g., as in the case ofbackplane-mounted daughterboards). In alternative embodiments, IC dice(i.e., chips) containing the transmitter 201 and receiver 209 may bepackaged within a single, multi-chip module with the chip-to-chipsignaling path formed by bond wires or other signal conductingstructures. Also, the transmitter 201 and receiver may be formed on thesame IC die (e.g., system on chip) and the signaling path 202implemented by a metal layer or other conducting structure of the die.

Referring to FIG. 4, the transmitter 201 transmits data on the signalingpath 202 during successive time intervals, referred to herein as symboltimes, T_(S). In the double-data-rate timing shown, each symbol timecorresponds to a half cycle of a transmit clock signal 208 (TCLK) suchthat two data values (e.g., values A and B) are transmitted on signalingpath 202 per cycle of the transmit clock signal 208. The transmitteddata signal arrives at the input of the receiver 209 after propagationtime, T_(P), and is sampled by the receiver 209 in response to edges ofa sampling clock signal 210 (SCLK). The sampling clock signal 210 may besupplied to the receive circuit 209 via an external clock line, or maybe a recovered version of a reference clock signal (e.g., recovered by adelay-locked loop or phase locked loop circuit). In other embodiments,discussed below, the sampling clock signal 210 may be recovered from theincoming data signal itself by a clock data recovery (CDR) circuit.Still referring to FIG. 4, the sampling clock signal 210 has aquadrature phase relation to data valid windows (i.e., data eyes) in theincoming data signal such that each sample of the incoming signal iscaptured at the midpoint of a data eye. In alternative embodiments, thesampling instant may be skewed relative to data eye midpoints asnecessary to satisfy signal setup and hold time requirements of thesamplers 211 and 213, and/or to compensate for asymmetry in the channelpulse response. Also, more or fewer symbols may be transmitted per cycleof the transmit clock signal 208. For example, the embodiment of FIG. 3may alternatively be a single data rate system, quad data rate system,octal data rate system, decade data rate system, and so forth.

In the receive circuit 209, a single symbol is captured during eachcycle of the sampling clock signal 210. That is, a rising (or falling)edge of the sample clock is used to capture a sample of the incomingsignal, x′_(n). In a multi-data rate system, multiple symbols arecaptured per cycle of the sampling clock signal 210 as shown in FIG. 4.In such systems, clock generation circuitry may be provided within thereceive-side device (e.g., an IC device containing the receiver 209) togenerate multiple instances of the sampling clock signal 210 that arephase-distributed through a period (1/frequency) of the sampling clocksignal. In the double-data-rate timing arrangement of FIG. 4, forexample, two instances of the sampling clock signal 210 are provided: aneven-phase sampling clock signal, SCLK_(E), to sample even-numberedsymbols x′_(n), x′_(n+2), x′_(n+4) . . . ; and an odd-phase samplingclock signal, SCLK_(O), to sample odd-numbered symbols x′_(n−1),x′_(n+1), x′_(n+3) . . . This technique may be extended to achievevirtually any data rate, including quad data rate (4 symbols persampling clock cycle), octal data rate (8 symbols per sampling clockcycle), decade data rate (10 symbols per sampling clock cycle), and soforth.

Still referring to FIG. 3, the transmitter 201 includes a transmit shiftregister 203, output driver bank 204 and tap weight register 206. In theparticular embodiment shown, the transmit shift register 203 is fiveelements deep and used to store a pre-tap data value D₊₁, primary datavalue D₀, and three post-tap data values D⁻¹, D⁻² and D⁻³. The primarydata value is the data value to be transmitted to the receiver 209during a given transmit interval, and the pre- and post-tap data valuesare the next-to-be transmitted and previously transmitted data values,respectively (i.e., the subscript indicating the number of transmitintervals to transpire before the data value will be transmitted). Eachof the shift register storage elements is coupled to a respective one ofoutput drivers 205 ₀–205 ₄ within the output driver bank 204, withoutput driver 205 ₁ forming the primary data driver, output driver 205 ₀forming the pre-tap data driver and output drivers 205 ₂–205 ₄ formingthe post-tap data drivers (such drivers being referred to herein aspre-tap, primary and post-tap drivers, for brevity).

The tap weight register is used to store a set of drive strength values,W _(N), referred to herein as tap weights. As described below, the tapweights are iteratively updated, with each new set of tap weights beingdesignated by an incrementally higher subscript (i.e., N, N+1, N+2,etc.). Each tap weight of a given set, W_(N)(0)–W_(N)(4), is supplied toa respective one of the output drivers 205 ₀–205 ₄ to control the levelof the output signal generated by the output driver. In one embodiment,the signal path 202 is pulled up to a predetermined voltage level (e.g.,at or near supply voltage) by single-ended or double-ended terminationelements, and the output drivers 205 ₀–205 ₄ generate signals on thesignal path 202 by drawing a pull-down current, I_(PD) (i.e., dischargecurrent), in accordance with the corresponding tap weight and datavalue. As a specific example, in a binary signaling system, each outputdriver 205 ₀–205 ₄ draws a current according to the followingexpression:I _(PD)(i)=S(i)·[W _(N)(i)*I _(UNIT)]  (1),where ‘·’ denotes a logic AND operation, ‘*’ denotes multiplication,I_(UNIT) is a reference current, W_(N)(i) is the tap weight of thei^(th) output driver (i ranging from 0–4 in this example), and S(i) isthe sign of the output driver contribution. The individual currentsdrawn by the output drivers 205 ₀–205 ₄ are wire-summed (i.e., drawnfrom the same node) to form a total pull-down current, I_(TPD), andtherefore each contribute to the total output signal level in accordancewith the sign of the output driver contribution and the tap weight. Bythis arrangement, pre- and post-tap drivers are enabled to provideadditive and subtractive contributions to the output signal level, asnecessary to compensate for dispersion-type ISI.

It should be noted that the particular numbers of pre-tap and post-tapdrivers (and corresponding tap weights and shift register elements)shown in FIG. 3 and the figures that follow have been selected forpurposes of example only. In alternative embodiments, more or fewerpre-tap drivers and/or post-tap drivers may be provided, along with moreor fewer storage elements within shift register 203 and tap weightswithin tap weight register 206.

In one embodiment, each of the tap weights, W_(N)(0)–W_(N)(4) is adigital value having a sign component and magnitude component. The signcomponent of the tap weight (e.g., sign bit) is exclusive-NORed with thecorresponding transmit data value to generate the sign of the signalcontribution to be generated by the corresponding output driver 205. Theexclusive-NOR operation effectively multiplies the signs of the tapweight and transmit data value, yielding a logic ‘1’ (i.e., interpretedas a positive sign in one embodiment) if the signs of the tap weight andtransmit data value are the same, and a logic ‘0’ (i.e., negative sign)if the signs of the tap weight and transmit data value are different.The magnitude component of the tap weight is a multi-bit value used, forexample, to control a digital-to-analog converter (DAC) within theoutput driver. Thus, the expression (1) may be rewritten as follows:I _(PD)(i)=[D(i)/⊕sgn(W _(N)(i))]·[|W _(N)(i)|*I _(UNIT)]  (2),where ‘/⊕’ denotes an exclusive-NOR operation, D(i) is a data valuereceived from the transmit shift register, “sgn(W_(N)(i))” is the signof the i^(th) tap weight and |W_(N)(i)| is the magnitude of the i^(th)tap weight. By this arrangement, the sign of the signal contributiongenerated by the i^(th) output driver is positive (i.e., logic ‘1’) ifthe sign of the corresponding tap weight and source data value match,and negative otherwise. That is, if a logic ‘1’ is to be transmitted(i.e., positive data) and the tap weight is positive (indicated by alogic ‘1’ sign bit), the signal contribution is positive, therebyincreasing the signal level generated on signal path 202. The signalcontribution is also positive if a logic ‘0’ is to be transmitted (i.e.,negative data) and the tap weight is negative, the negative tap weighteffectively flipping the otherwise negative signal contributionindicated by the logic ‘0’ data. If the tap weight sign and source datavalue do not match, then a negative signal contribution is generated bythe output driver. In a multi-level signaling embodiment, the sign ofthe tap weight may similarly be used to change the sign of thetransmitted symbol.

FIG. 5 illustrates an embodiment of a differential output driver 230that may be used to implement each of the output drivers 205 ₀–2054 ofFIG. 3. The output driver includes a pair of transistors 233 and 235,each having drain terminals pulled up by respective load elements R(resistors are depicted in FIG. 5, but active load elements or othertypes of resistive elements may alternatively be used) and coupled tonegative and positive lines 240 and 242 (L− and L+ respectively) ofdifferential signal path 202. Source terminals of the transistors 233and 235 are coupled in common to a current DAC 237 (IDAC) which draws acurrent, I_(S), in accordance with the magnitude component of tapweight, W_(N)(i). That is, I_(S)=|W_(N)(i)|×I_(UNIT). An exclusive-NORgate 231 is provided to exclusive-NOR the sign of the tap weight 232with the corresponding source data value 234, thereby generating asignal contribution sign, S(i), that is supplied to the gate oftransistor 233. The complement of the signal contribution sign, /S(i) isgenerated by inverter 239 and supplied to the gate of transistor 235. Bythis arrangement, when a logic ‘1’ data value 234 is received in theoutput driver, and the tap weight 232 is positive, a positivecontribution sign is generated by the exclusive-NOR gate (i.e., S(i) ishigh) to switch on transistor 233 and switch off transistor 235, therebycausing line 242 (L−) to be pulled down relative to line 240 (L+) toestablish a positive differential signal contribution. The potentialdifference between lines L+ and L− is controlled by the current I_(S)(i.e., V_(L+)=V_(S−)I_(S)R, where the supply voltage, V_(S), and theresistance, R, are substantially fixed) which, in turn, is controlled bythe magnitude component of the tap weight 232. Thus, the signs of thetap weight 232 and source data value 234 control whether thedifferential signal contribution generated on lines 240 and 242 by agiven output driver (i.e., V_(L+)−V_(L−)) is positive or negative, andthe magnitude of the tap weight 232 controls the amplitude of thedifferential signal. In alternative embodiments, described in greaterdetail below, multi-level signaling (i.e., signaling protocols in whicheach transmitted symbol carries more than one bit of information) may beused instead of binary signaling, with different pull down currentsbeing used to establish different signal levels for differentcombinations of source data bits. Also push-pull type output drivers orother types of output drivers may be used instead of the current modedriver 230 shown in FIG. 5.

Output Driver Tap Weight Determination

Referring again to FIG. 3, the tap weights stored in tap weight register206 are ideally set to exactly cancel the dispersion-type ISI (and/orother systematic sources of signal distortion) resulting fromtransmission of the pre-tap and post-tap data values. For example, iftransmission of a logic ‘1’ value that starts at a normalized signallevel of +1.0 results in reception of signals having levels of 0.7 and0.3 in successive reception intervals, then the signal is beingdispersed and attenuated by the signal path 202 (also referred to hereinas a channel). Consequently, an immediately subsequent transmission of alogic ‘0’ that starts at a normalized signal level of −1.0 results inreception of signals having levels of −0.4 (i.e., −0.7+0.3), and −0.3.That is, the residue of the initial transmission (i.e., 0.3)destructively combines (i.e., interferes) with the subsequentnegative-level signal, attenuating the received signal level. In thissimple example, it can be seen that the source of the ISI in any givensymbol transmission is the immediately preceding symbol. Thus, bysetting the post-tap driver 205 ₂ to generate a subtractive pre-emphasissignal that exactly cancels the residue of the preceding transmission,the signal received within a given sampling interval, while not fullyrestored to the originally transmitted level, is free from ISI. In apractical application, the ISI will not be fully canceled, as numerousother channel effects (reflections, cross-talk, noise) mask the truelevel of ISI at any given time, making it difficult to ascertain theexact tap weight that should be applied to the pre- and post-tap drivers(i.e., 205 ₀ and 205 ₂–205 ₄) to compensate for the pre- and post-tapresidue. Also, the pre-emphasis signal itself will generate ISI, whichin turn may be mitigated by additional pre-emphasis signals generated byone or more others of the pre- and post-tap drivers.

In one embodiment, the receiver 209 generates updated tap weights, W_(N+1), based upon a comparison of incoming signals with an adaptivelydetermined target signal level 220, referred to herein as a data levelthreshold, DLEV. The receiver 209 includes an adaptive module 215 (AM)and a pair of sampling circuits referred to herein as a data sampler 211(D) and an adaptive sampler 213 (A). The data sampler samples theincoming signal, referred to herein as x′_(n) to emphasize the channeltransformation of originally transmitted signal, x_(n), and generates adata sample 216 (RX Data) having a logic ‘1’ or logic ‘0’ stateaccording to whether the incoming signal exceeds a zero reference. In asingle-ended signaling system, the zero reference may be generated by aDAC, voltage divider or other circuit and set to a point midway betweensteady-state high and steady-state low signaling levels. In adifferential signaling system, the common mode of the incomingdifferential signal may constitute the zero reference so that if thesignal level on the positive signal line (e.g., line 240 of FIG. 5)exceeds the signal level on the negative signal line (e.g., line 242 ofFIG. 5), a logic ‘1’ is captured by the data sampler 211 and,conversely, if the signal level on the negative signal line exceeds thesignal level on the positive signal line, a logic ‘0’ is captured by thesampler 211. Thus, the data sample 216 has a logic state thatcorresponds to the sign of the incoming data signal, positive ornegative, and is referred to herein as a data sign value.

The adaptive sampler 213 also samples the incoming signal, x′_(n), andgenerates an error sample 218 having a logic ‘1’ or logic ‘0’ stateaccording to whether the incoming signal exceeds the data levelthreshold 220 generated by the adaptive module 215. In one embodiment,the data level threshold 220 corresponds to an expected data level oflogic ‘1’ transmission, so that if the incoming signal is determined tohave a positive sign (i.e., RX Data=sgn(x′_(n))=‘1’), then the errorsample 218 generated by the adaptive sampler 213 represents the sign ofan error between the incoming signal level and the expected signal level(i.e., the data level threshold 220, DLEV). Accordingly, the errorsample 218 is referred to herein as an error sign value (sgn(e_(n))) andis a logic ‘1’ (i.e., positive) if x′_(n)<DLEV, and a logic ‘0’ (i.e.,negative) if x′_(n)≧DLEV).

The adaptive module 215 receives the data sign and error sign values,216 and 218, from the data sampler 211 and adaptive sampler 213,respectively, and adaptively updates the data level threshold 220 andpre-emphasis tap weights 226 in response. Referring to FIG. 6, byupdating the data level threshold 220 and tap weights 226 concurrently(i.e., at least partly overlapping in time at the same or different loopupdate rates), and by maintaining the updated tap weights 226 in anaggregate setting that corresponds to the peak (or average) power of thetransmit circuit 201, the data level threshold converges to theattenuated levels, +/−a exhibited by the highest frequency data patternstransmitted over the signal path 202, and the tap weights 226 convergeto a setting that substantially flattens the channel response as shownat 247. That is, instead of attempting to adapt the tap weightsaccording to originally transmitted signal levels (e.g., normalized +/−1levels as discussed in reference to FIGS. 1 and 2), the attenuatedsignal levels +/−a of high frequency data patterns (e.g., signal levelsalternating in each successive transmission) are learned and used togenerate error signals that, when applied in an error reduction circuit,drive the pre-emphasis tap weights 226 toward a solution that flattensthe channel response at the +/−a threshold levels, and yet meets thepeak power constraint of the transmit circuit. By this operation, a moreoptimal tap weight convergence may be achieved than in the prior-artsignaling system of FIG. 1, potentially improving signaling margins,particularly in multi-PAM systems where finer distinctions betweensignaling levels are needed.

In some systems, it is desirable to shape the frequency responsedifferently from the flattened response described in reference to FIG.6. This may be done, for example, by updating both the tap weights anddata level threshold using the error filtered by appropriate datasequences. As an example, in a system where it is desired to pass theadditive or subtractive component (i.e., partial response) of aneighboring symbol that appears in the same transmit interval as thesymbol of interest (i.e., not zeroing the ISI from a selectedneighboring symbol). This data filtering, however, does not change theoperations described above with regard to concurrent updating of boththe data level threshold and tap weights (with or without powerscaling). Rather, the target shape of the pulse is changed. In otherembodiments, the tap weights updated using other error filteringfunctions to improve any number of performance measures (e.g., eyeopening in voltage or timing, reduced bit error rate or other overallsystem performance parameter).

In one embodiment, each new set of updated tap weights 226 iscommunicated to the transmitter via a back channel 225. The back channel225 may be formed, for example, by a relatively low-speed signalingpath, or by out-of-band signaling over the signaling path 202 (e.g.,using an otherwise unused code space within a signal encoding protocolsuch as 8b/10b or other signal encoding). In an alternative embodiment,a separate back channel may be omitted and the signaling path 202 may beused to communicate updated tap weights 226 (or update values thatenable transmit-side generate of updated tap weights 226) to thetransmit-side device.

Adaptive Module

FIG. 7 illustrates an embodiment of an adaptive module 250 that may beused to implement the adaptive module 215 of FIG. 3. The adaptive module250 includes a data sign register 251, error sign register 253, signmultiplier 257, finite state machine 255, power scaling logic 259,filter 261, threshold counter 269 and DAC 271. The error sign value 218and data sign value 216 generated during reception interval ‘n’ aresupplied to the error sign register 253 and data sign register 251,respectively, and clocked into the registers in response to transitionsof a sampling clock signal, not shown (or other, related clock signal).The data sign register 251 is a shift register used to store the mostrecently generated data sign values. In the embodiment of FIG. 7, thedata sign register 251 is depicted as being five elements deep (i.e., tostore data sign values, x′_(n−1)–x′_(n−4)); a depth that corresponds tothe number of tap weights applied within the transmit circuit 201 ofFIG. 3. In alternative embodiments, the data sign register 251 may havemore or fewer storage elements, for example, to accommodate more orfewer tap weights and/or to store data sign values used for otherpurposes including, without limitation, reflection cancellation,cross-talk cancellation and offset cancellation. Similarly, the errorsign register 253 is a one-deep register in the embodiment of FIG. 6, tostore error sign value sgn(e_(n−1)), but may include any number of shiftregister elements in alternative embodiments (e.g., to enable selectionof an error sign value having a desired latency).

The sign multiplier 257 includes a set of exclusive-NOR gates 258 ₀–258₄ each having a first input coupled in common to receive the storederror sign value from the error sign register 253 and each having asecond input coupled to receive a respective data sign value from thedata sign register 251. By this arrangement, each of the exclusive-NORgates 258 ₀–258 ₄ generates a respective one of update values 260,UD(0)–UD(4), in a logic ‘1’ state if the corresponding data sign valuematches the error sign value, and in a logic ‘0’ state if the data signvalue and error sign value do not match. Thus, each of the update values260 represents a multiplication of the signs of the input signal (i.e.,x′_(n−1)–x_(n−4), respectively) and error signal e_(n−1), and thereforeis a logic ‘1’ if the signs are both positive or both negative, and alogic ‘0’ if the signs are different. In one embodiment, each of theupdate values 260 is filtered within a respective one of filter elements262 (F) to decrease update dither due to noise in the update estimate.In an alternative embodiment, the filter elements 262 are omitted.

In one embodiment, a tap weight is made more positive in response to alogic ‘1’ update (i.e., a positive update) and more negative in responseto a logic ‘0’ update value (a negative update). More specifically, apositive tap weight is incremented (e.g., by a predetermined step size)and a negative tap weight decremented in a positive update. Conversely,a positive tap weight is decremented and a negative tap weightincremented in a negative update. In one embodiment, the positive andnegative updates applied to the tap weights constitute a sign-signleast-mean-square (LMS) update that may be expressed as follows:W _(N+1) =W _(N)+stepsize*sign(e _(n))*sign( x ′)   (3),which corresponds to the following scalar expressions:W _(N+1)(0)=W _(N)(0)+stepsize*sign(e _(n))*sign(x′ _(n+1))W _(N+1)(1)=W _(N)(1)+stepsize*sign(e _(n))*sign(x′ _(n))W _(N+1)(2)=W _(N)(2)+stepsize*sign(e _(n))*sign(x′ _(n−1))W _(N+1)(3)=W _(N)(3)+stepsize*sign(e _(n))*sign(x′ _(n−2))W _(N+1)(4)=W _(N)(4)+stepsize*sign(e _(n))*sign(x′ _(n−3))Thus, each tap weight update is in the direction of the estimate of thequantized negative gradient of the quadratic, least-mean-squared errorcost function (i.e., a quadratic cost function). Other cost functionsmay be used in alternative embodiments. In order to provide a meaningfulerror signal, the data level threshold is updated according to the errorsign value, sign(e_(n)). In the embodiment of FIG. 3, for example, thedata level threshold is updated according to the following expression:DLEV _(N+1) =DLEV _(N)−stepsize*sign(e _(n))*sign(x′_(n))

In the embodiment of FIG. 7, the adaptive module 250 outputs the updatedtap weight values 226 generated by the power scaling logic 259 to thetransmit-side device, for example, via the back channel 225 depicted inFIG. 3 (or via another signaling path). In an alternative embodiment,the power scaling logic 259 is provided within the transmit-side devicerather than the receive-side device, so that only the tap weight updates(or component signals used to generate the tap weight updates) need becommunicated to the transmit-side device.

Still referring to FIG. 7, the most recently stored data sign value anderror sign value, sgn(x′_(n)) and sgn(e_(n−1)), are provided to thefinite state machine 255 which, in turn, asserts an update-weight signal282 (UW) to enable the power scaling logic 259 to apply the updatevalues 260 to the existing set of tap weights (W _(N)), and scale theresulting values to generate updated tap weights W _(N+1) 226. In theembodiment of FIG. 7, the finite state machine asserts the update-weightsignal upon determining that the shift register 251 is fully loaded, orfully reloaded, with a set of data sign values, and that the mostrecently stored data sign value has a predetermined state. Thepredetermined state may be either positive or negative in differentembodiments, according to whether the data level threshold 220 generatedby the adaptive module 250 corresponds to positive or negative incomingsignals. That is, if the data level threshold 220 is adjusted to thelevel of logic ‘1’ data, then the error signal, e_(n), has meaning withrespect to x′_(n) if the sign of x′_(n) is positive (i.e., the data signvalue is a logic ‘1’) and is ignored if the sign of x′_(n) is negative.Conversely, if the data level threshold 220 is adjusted to the level oflogic ‘0’ data, then the error signal, e_(n), has meaning with respectto x′_(n) if the sign of x′_(n) is negative and is ignored if the signof x′_(n) is positive. Further, two adaptive samplers may be provided togenerate positive and negative data level thresholds when positive andnegative data signals are received, respectively. As discussed below, ina multi-PAM embodiment, an adaptive sampler may be provided to generateerror information for each different data level.

In the embodiment of FIG. 7, the adaptive module 250 generates a datalevel threshold 220 (DLEV) that constitutes a target data level forincoming, positive data signals. When the finite state machine 255detects storage of a positive data sign value (i.e., a logic ‘1’), thefinite state machine 255 asserts an update threshold signal 268 (UT),thereby enabling a threshold count 270 maintained by threshold counter269 to be incremented or decremented according to the state of thecorresponding error sign value, e_(n−1), stored in register 253. Filter267 is provided to decrease update dither due to noise in the updateestimate, and may be omitted in alternative embodiments. Also, thefinite state machine 255 may also generate the update threshold signal268, upon determining that a predetermined pattern of incoming signalshas been received (e.g., a high-frequency pattern such as 10101).

In the embodiment of FIG. 7, the threshold counter 269 outputs thethreshold count 270 to a DAC 271 which, in turn, generates acorresponding data level threshold 220. Although depicted as being partof the adaptive module 250, the DAC may alternatively be a componentwithin the adaptive sampler 213 (e.g., a DAC that operates to bias thesampler to establish the data level threshold). In such an embodiment, adigital control value (i.e., the threshold count 270) is output from theadaptive module 250 rather than an analog threshold level (or analogbiasing signal). Sampling circuit embodiments having biasing circuitryto establish a data level threshold in response to a digital controlvalue are described below.

Still referring to FIG. 7, the adaptive module may additionally includea filter 261, offset counter 263 and DAC 265 to control offsetcancellation within the data sampler. During an offset cancellationoperation, an offset adjust signal 252 is asserted at an input of thefinite state machine, and a null signal is generated at the data samplerinput, for example, by switchably coupling the sampler inputs together,or by transmitting null data over the signal path (i.e., signal levelsimpressed on component lines of the differential signal path have thesame levels). A steady-state positive or negative output from the datasampler in response to the null data input indicates a DC error withinthe sampler. That is, if the sampler repeatably interprets nominallyequal signal levels at its differential inputs as indicating a logic ‘1’or logic ‘0’ value, then the sampler exhibits a DC offset. Accordingly,the data sign value, after being filtered by the filter 261 (which maybe omitted in alternative embodiments), is supplied to an up/down inputof the offset counter 263. The finite state machine responds toassertion of the offset adjust signal by asserting an update-offsetsignal 284 (UO) after each new data sign value is loaded into the shiftregister (or after a predetermined number of data sign values have beenloaded), thereby enabling the offset count 264 maintained within theoffset counter 263 to be adjusted up or down. In the embodiment of FIG.7, the offset count 264 is supplied to DAC 265 which, in turn, generatesan analog control value 266 (OFST) that is applied within the datasampler to bias the sampler in a direction counter to the DC offset.Alternatively, the offset count 264 itself may be supplied to thesampler. In either case, a negative feedback loop is created in whichthe data sampler bias is adjusted to drive the DC offset to zero, acondition indicated by a dithering offset count 264. In one embodiment,the offset count 264 is supplied to the finite state machine 255 (orother control circuit) to enable the finite state machine 255 todetermine when a target DC offset count has been reached (i.e., offsetcalibration operation complete). In alternative embodiments, the finitestate machine 255 continues to assert the update-offset signal 284(i.e., continuing the DC offset calibration operation) until the offsetadjust signal 252 is deasserted. The offset adjust signal 252 may beasserted, for example and without limitation, for a predetermined time,or until a predetermined number of data sign values have been generated,or until a dithering offset count is detected.

In one embodiment, the offset count 264 (or DAC output 266) is suppliedto both the adaptive sampler and the data sampler (e.g., elements 213and 211 of FIG. 3), on the assumption that the DC offset of the adaptivesampler is likely to track the DC offset of the data sampler. This maybe the case, for example, when a contributor to DC offset is the signalpath itself, or when the DC offset is process dependent. In analternative embodiment, additional offset calibration circuitry (e.g.,filter, offset counter and, if needed, DAC) is provided within theadaptive module 250 to enable DC offset calibration of the adaptivesampler. In another alternative embodiment, multiplexing circuitry isused to select the error sign register 253 to provide the sample valueto the filter 261 instead of the data sign register 251. In suchalternative embodiments, the threshold count applied to the adaptivesampler is temporarily zeroed (or disabled from being applied within theadaptive sampler) to enable determination of the DC offset.

Power Scaling

Still referring to FIG. 7, after the power scaling logic 259 (or othercircuitry within the adaptive module) updates the transmit pre-emphasistap weights according to the update values, the power scaling logic 259scales the updated tap weights to ensure that the total power indicatedby the aggregate magnitudes of the tap weights does not exceed the powerconstraint (peak or average) of the transmit circuit. In one embodiment,the power constraint of the transmit circuit corresponds to the maximumDAC setting of the primary driver which, in an 8-bit sign-magnitudeimplementation, is 2⁷−1=127 (alternatively, the maximum DAC setting, andtherefore the power constraint, may be programmed into a configurationcircuit within the receive-side and/or transmit-side device, or suppliedto the receive-side and/or transmit-side devices during systeminitialization). Thus, assuming an initial condition in which theprimary driver tap weight is set to max power (i.e., sign bit=1,magnitude=127), then as the magnitudes of the initially-zero pre- andpost-tap weights increase, the power constraint may be exceeded. Asdiscussed above, the sign-sign LMS update logic of FIG. 7 updates thetap weights according to the following equation:W _(N+1) =W _(N)+stepsize*sign(e _(n))*sign( x ′)   (3).Thus, the tap weight updates are obtained by multiplying the stepsize,error sign value and data sign value, so that expression (3) may berewritten as follows:W _(N+1) =W _(N)+Update _(N)   (4).The transmit circuit power constraint may be expressed as a sum of themagnitudes of the output driver tap weights. That is:

-   Σ|Wn|<=W_(MAX), where W_(MAX) is the square root of the normalized    power limitation (i.e., in the case of a peak power constraint; in    the case of an average power constraint, the expression becomes the    L2 norm: ΣWn²<=W_(MAX) ²). In a current mode transmitter, the tap    weights, W, control the current contribution of each output driver,    which in turn controls the voltage level developed on the signaling    path and therefore the power output of the drivers. In a voltage    mode transmitter, the tap weights control the voltage contribution    of each output driver, and therefore the power output of the    drivers. In the tap weight update expressions herein, the term,    W_(MAX), refers to the square root of the normalized peak or average    power constraint.    In one embodiment, transmit pre-emphasis tap weights are re-scaled    directly after each update by multiplying each tap weight magnitude    by a ratio of the power constraint to the power represented by the    updated tap weights. That is:    W _(N+1)=( W _(N)+Update _(N))*(W _(MAX) /|W _(N)+Update _(N)|₁)      (5),    where |W _(N)+Update _(N)|₁ is the sum of the magnitudes of the tap    weights that would result if the updates were applied (i.e.,    |W_(N)(0)+Update(0)|+|W_(N)(1)+Update(1)|+ . . .    +|W_(N)(4)+Update(4)|). Direct re-scaling may be carried out by a    processing unit (e.g., digital signal processor, special purposes    processor, or general purposes processor) within either the    receive-side IC device or transmit-side IC device (i.e., the IC    devices that include the receiver 209 and transmitter 201,    respectively, of FIG. 3) or by another device. Alternatively, a    state machine or dedicated logic circuit for carrying out the direct    re-scaling operation (e.g., using integer arithmetic) may also be    used.

In an alternative embodiment, circuitry within the adaptive moduleitself is used to carry out re-scaling based on a Taylor-seriesapproximation that reduces computational complexity relative to thedirect re-scaling approach. That is, rewriting expression (4), thefollowing expression for residual power (i.e., amount of power by whichthe updated tap weights exceed or fall below the power constraint) isobtained:W _(RES) =|W _(N)+Update _(N)|₁ −W _(MAX) =Σ[sgn(W_(N)(i))*Update_(N)(i)]  (6).Combining expressions (5) and (6), the direct re-scaling operation maybe expressed as a ratio of the residual power and the power limit:W _(N+1)=( W _(N)+Update _(N))*[1+W _(RES) /W _(MAX)]⁻¹  (7).Using the Taylor-series approximation, [1+W_(RES)/W_(MAX])⁻¹≈[1−W_(RES)/W_(MAX)], expression (7) may be rewritten as follows:W _(N+1)≈( W _(N)+Update _(N))−[( W _(N)+Update _(N))*W _(RES) /W_(MAX)]  (8).Expression (8) may be implemented in a relatively small logic circuitconsidering that the term (W _(N)+Update _(N)) may be obtained throughinteger addition, and, because W_(RES) will usually be significantlysmaller than W_(MAX), the multiplication by W_(RES)/W_(MAX) can bereduced to a right-shift, binary division operation. That is, 1/W_(MAX)involves a right shift by log2(W_(MAX)) bits, so long as W_(MAX) is apower-of-two value (e.g., 128). Similarly, W_(RES), which ranges from +5to −5 in the five-driver embodiment of FIGS. 3 and 7, will be a power of2 value in all cases except for +/−3 or +/−5, which may be rounded to apower of 2 number. In one embodiment, for example, +/−3 W_(RES) valuesare alternately rounded to +/−2 and +/−4. W_(RES) values of +/−5 arerounded to +/−4. Different rounding schemes may be used in alternativeembodiments. For example, W_(RES) values of +/−5 may be rounded bytoggling between 8 and 4 (e.g., rounding to 8 once for every threeroundings to 4).

FIG. 8 illustrates a residue-based power scaling circuit 290 thatoutputs scaled, updated tap weights in accordance with the approximationset forth in expression (8). The power scaling circuit 290 includes abank of exclusive-NOR gates 291 ₀–291 ₄ that multiply the signs of theexisting tap weights (W _(N)) stored in registers 302 ₀–302 ₄ with thesigns of the update values 260 (i.e., UD(0)–UD(4)). A summation circuit293 receives the outputs of the exclusive-NOR gates 291 and generates asum that corresponds to the residual power (W_(RES)). That is, thesummation circuit treats each logic ‘1’ input as a +1 value and eachlogic ‘0’ value as a −1 value, thereby generating a residual power value294 that indicates the aggregate change in tap weights. In theembodiment of FIG. 8, the residual power value 294 is a sign-magnitudevalue having a sign component 310 (i.e., sign bit) that indicateswhether the aggregate change in tap weights is positive or negative, anda magnitude component 297 that represents the absolute value of theaggregate change in tap weights. The magnitude component 297 of theresidual power value 294 is input to a shift control circuit 295 that,in turn, generates a shift value 298 (S#), which corresponds to thenumber of bits by which an updated tap weight is to be right shifted tocarry out a multiplication by |W_(RES)|/W_(MAX). That is, the shiftvalue 298 corresponds to log₂(W_(MAX)/|W_(RES)|). In the embodiment ofFIG. 8, the maximum power is assumed to be 128 so that, as shown inlogic table 296, the shift control circuit 295 generates a shift value298 of eight when the residual power value 294 is zero; a shift value ofseven when the residual power value is one; a shift value of six whenthe residual power is two; alternating shift values of five and six whenthe residual power value is three; and a shift value of five when theresidual power value is greater than three.

The update values 260 and existing tap weights in registers 302 are alsosupplied to respective scaling circuits 301 ₀–301 ₄ along with the shiftvalue 298, and the sign component 310 of the residual power value 294.Referring to the detailed view of scaling circuit 301 ₄, the updatevalue, UD(4), and tap weight W_(N)(4) are input to anincrement/decrement circuit 303 which generates an updated tap weightvalue 304 having an incremented magnitude if the tap weight and updatehave the same sign (i.e., both positive or both negative) and adecremented magnitude if the tap weight and update have different signs.In the embodiment of FIG. 8, the updated tap weight value 304 includes asign component 312 which is supplied to the first input of anexclusive-OR gate 307, and a magnitude component 306 which is suppliedto a shifting circuit 305 (e.g., a barrel shifter). The second input ofthe exclusive-OR gate 307 is coupled to receive the sign component 310of the residual power value 294 SO that the exclusive-OR gate outputs alogic ‘1’ select signal 314 to the select input of multiplexer 315 ifthe sign of the updated tap weight value and the sign component of theresidual power are different, and a logic ‘0’ select signal 314 if thesign components of the updated tap weight value and residual power arethe same. The complete updated tap weight value 304 (i.e., sign andmagnitude) is provided to difference circuit 309 and summing circuit311. The shifting circuit 305 right shifts the magnitude component 306of the updated tap weight 304 according to the shift value 298 toeffectuate a multiply by W_(RES)/W_(MAX) (or an approximation ofW_(RES)/W_(MAX)) and outputs the resulting product to the summingcircuit 311 and difference circuit 309. The summing circuit 311 adds theproduct generated by the shifting circuit 305 to the updated tap weightvalue 304 and, the difference circuit 309 subtracts the productgenerated by the shifting circuit 305 from the updated tap weight 304 togenerate scaled-up and scaled-down updated tap weight values,respectively, which are provided, in turn, to first and second inputports of the multiplexer 315. By this arrangement, if the signcomponents 312 and 310 of the updated tap weight value 304 and residualpower value 294, respectively, are the same, then the scaled-downupdated tap weight value generated by the difference circuit 309 isselected by multiplexer 315 to be output as the updated tap weight 308 ₄(i.e., ultimately to become updated tap weight W_(N+1)(4)). If the signcomponents 312 and 310 of the updated tap weight value 304 and residualpower value 294, respectively, are different, then the scaled-up updatedtap weight value generated by the summing circuit 311 is selected bymultiplexer 315 to be output as the updated tap weight 308 ₄. Thus, inthe case of a positive residual power value 294, a positive tap weightvalue is scaled down and a negative tap weight value is scaled up (i.e.,made less negative) to reduce the power applied within the correspondingoutput driver. Conversely, in the case of a negative residual powervalue 294, a negative tap weight value is scaled down (i.e., made morenegative) and a positive tap weight value is scaled up to increase thepower applied within the corresponding output driver. Thus, each ofupdated tap weights W_(N+1)(0)–W_(N+1)(4) is generated within arespective one of scaling circuits 301 ₀–301 ₄ by adjusting the priortap weight (W_(N)), multiplying the adjusted tap weight by theW_(RES)/W_(MAX) approximation to generate a fractional component (i.e.,the output of shifting circuit 305), then subtracting the fractionalcomponent from the updated tap weight (note that an addition occurs whena negative W_(RES) is subtracted from the updated tap weight). That is,W _(N+1) is assigned the value: (W _(N)+Update _(N))−[(W _(N)+Update_(N))*W_(RES)/W_(MAX)], the Taylor-series approximation set forth abovein expression (8). In one embodiment, the updated tap weights 308 ₀–308₄ are stored within the registers 302 ₀–302 ₄ in response to assertionof the update-weight signal 282 (UW). Alternatively, the update-weightsignal 282 is used to initiate operation of a finite state machine (orother logic circuit) which controls and times the increment, shift andsubtract operations within the scaling circuits 301 and other logiccircuits within the power scaling logic 290, culminating in storage ofthe updated tap weights 308 in registers 302. In either case, oncestored, the updated tap weight values 308 become the existing tap weightvalues 226 that are supplied to the exclusive-NOR gates 291 and scalingcircuits 301 to generate the next set of updated tap weights 308.

FIG. 9 illustrates an alternative embodiment of a power scaling logiccircuit 320 referred to herein as a power bounding embodiment. In thepower bounding embodiment, tap weight updates 260 are applied to adjustthe pre-emphasis tap weights first (i.e., the tap weights applied to thepre- and post-tap drivers), then the magnitudes of the adjustedpre-emphasis tap weights and the primary driver tap weight are summed togenerate an aggregate magnitude. The magnitude of the primary driver tapweight (i.e., the primary tap weight) is then decreased or increased ifthe aggregate magnitude exceeds the power constraint or falls below apredetermined lower bound, respectively. By this operation the totalpower applied to the transmit circuit output drivers is maintainedbetween an upper and lower bound.

The power scaling logic 320 includes a set of tap weight counters 325₀–325 ₄, state counter 324, operand multiplexer 327, accumulator 329,and primary update logic 341. Tap weight updates 260 for the pre- andpost-tap driver tap weights are supplied to the power scaling logic 320along with negative versions of upper and lower power bound values, 323and 321, respectively, and the update-weight signal 282. At the start ofan update event, the update-weight signal 282 is asserted to enable thetap weight counters for the pre- and post-tap weights (i.e., 325 ₀ and325 ₂–325 ₄) to be incremented or decremented according to the state ofthe corresponding update signal 260. Assertion of the update weightsignal also triggers the state counter 324 to roll over from a finalstate count of seven, to an initial state count of zero and enables thestate counter 324 to auto increment from zero to seven. The state count326 is supplied to the operand multiplexer 327 so that, as the statecount 326 progresses from zero to six, the operand multiplexer 327outputs, in turn, the magnitudes of the updated tap weights stored incounters 325 ₀, 325 ₂, 325 ₃ and 325 ₄, and the negative lower and upperpower bounds, 321 and 323, to the accumulator 329.

The accumulator 329 includes a temporary register 333 (TREG), summingcircuit 335 and multiplexer 331. The multiplexer 331 has a control inputcoupled to receive the state count 326 and three input ports coupledrespectively to the outputs of the operand multiplexer 327, summingcircuit 335 and temporary register 333. When the state count 326 iszero, the multiplexer 331 outputs the operand 330 selected by theoperand multiplexer 327 (i.e., the magnitude of updated pre-tap weight,|W_(N+1)(0)|, maintained within tap weight counter 325 ₀); when thestate count 326 is one, two, three or four, the multiplexer 331 outputsthe sum generated by the summing circuit, and when the state count 326is five and above, the multiplexer 331 outputs the content of thetemporary register. The summing circuit 335 has first and second inputscoupled respectively to the outputs of the operand multiplexer 327 andthe temporary register 333. The temporary register 333 is coupled toreceive the output of the multiplexer 331 and is re-loaded in responseto each transition of the state count 326. By this arrangement, when thestate count 326 is zero, the magnitude of the updated pre-tap weight,|W_(N+1)(0)| is applied to the input of the temporary register 333. Whenthe state count 326 transitions from zero to one, the temporary register333 is loaded with the magnitude of the pre-tap weight, and themagnitude of the primary tap weight, |W_(N)(1)| is output by the operandmultiplexer 327 and summed with the magnitude of the pre-tap weight(i.e., the content of the temporary register 333) in summing circuit335. The sum of tap weight magnitudes W_(N+1)(0) and W_(N)(1) isselected by the multiplexer 331 (i.e., in response to state count=1) andsupplied to the input of the temporary register. Accordingly, when thestate count 326 transitions from one to two, the sum of tap weightmagnitudes |W_(N+1)(0)| and |W_(N)(1)| is loaded into the temporaryregister 333 and supplied to the summing circuit 335 for summation withthe magnitude of the updated post-tap weight, |W_(N+1)(2)| (i.e., thetap weight magnitude selected by the operand multiplexer 327 in responseto state count=2). By this operation, as the state count 326 isincremented from zero to four, a sum of the tap weight magnitudes isaccumulated in the temporary register 333, culminating in storage of thesum of the magnitudes of all the tap weights (i.e.,|W_(N+1)(0)|+|W_(N)(1)|+|W_(N+1)(2)|+|W_(N+1)(3)|+|W_(N+1)(4)|) withinthe temporary register 333 when the state count 326 transitions fromfour to five. The sum of magnitudes of all the tap weights representsthe power in the updated tap weights, prior to updating the primary tapweight and is referred to herein as a proposed power value. When thestate count 326 is five and above, the multiplexer 331 selects theoutput of the temporary register to be re-loaded into the temporaryregister, effectively placing the temporary register 333 in a hold stateto maintain the proposed power value therein. In an alternativeembodiment, the temporary register 333 is not re-loaded after the countvalue reaches 5, thereby maintaining the proposed power value in thetemporary register 333.

Still referring to FIG. 9, when the state count 326 reaches five, theoperand multiplexer 327 outputs the negative lower power bound 321 tothe accumulator 329 which, by operation of summing circuit 335,subtracts the lower power bound value 321 from the proposed power value.The sign of the difference between the proposed power value and lowerpower bound value 321 constitutes a lower-bound comparison result (LBC)that indicates whether the proposed power value is greater than (orequal to) the lower power bound value (i.e., LBC=0) or less than thelower power bound (LBC=1) and is supplied to the primary update logic341. The primary update logic includes a storage element 343 (e.g., a Dflip-flop as shown in FIG. 9, a latch or other storage circuit),exclusive-NOR gate 345 and logic AND gate 347. As the state count 326transitions from five to six, the lower-bound comparison result 338 isstored in the storage element 343 and is output therefrom as a storedlower bound compare result 338 until the next five-to-six state counttransition. Also, the operand multiplexer 327 selects the negative upperpower bound value 323 to be summed with the proposed power value insumming circuit 335, effectively subtracting the upper power bound value323 from the proposed power value. The sign of the difference betweenthe proposed power value and the upper power bound value 323 constitutesan upper-bound comparison result 336 (UBC) that indicates whether theupper power bound is greater than the proposed power value (i.e.,sign=1) or less than or equal to the proposed power value (i.e.,sign=0). Thus, as the state count transitions from six to seven, theupper- and lower-bound compare results 336 and 338 indicate equalities(and inequalities) adjustments to the primary tap weight, as shown inthe following table (PP=Proposed Power, UB=Upper Bound, LB=Lower Bound,PTW=Primary Tap Weight):

TABLE 1 UBC LBC Equality Indication Update PTW? PTW Adjustment 0 0 PP ≧UB 1 (Yes) Decrement PTW 0 1 Invalid (PP ≧ UB & 0 (No) — PP < LB) 1 0UB > PP ≧ LB 0 No Adjustment 1 1 PP < LB 1 Increment PTW

Still referring to FIG. 9, the upper-bound compare result 336 and storedlower-bound compare result 338 are supplied to respective inputs of theexclusive-NOR gate 345 to generate a primary tap weight update signal346 in accordance with Table 1. The AND gate 347 receives the primarytap weight update signal 346 at a first input and an indication that thestate count has reached seven at a second input. By this arrangement, asthe state count transitions from six to seven, the AND gate 347 assertsan update enable signal 348 if the lower- and upper-bound compareresults have the same state (i.e., either both ‘1’s or both ‘0’s). Theupdate enable signal 348 is supplied to a count enable input (i.e.,strobe input) of the primary tap weight counter 325 ₁, and theupper-bound compare result 336 is supplied to an up/down input of thecounter 325 ₁. Consequently, if the update enable signal is asserted,the primary tap weight is incremented in response to a logic ‘1’upper-bound compare result 336 (i.e., indicating that both UBC and LBCare high and therefore that the proposed power is below the lower bound)and decremented in response to a logic ‘0’ upper-bound compare result336 (i.e., indicating that both UBC and LBC are low and therefore thatthe proposed power is above or equal to the upper power bound 323). Notethat the upper power bound value 323 input to the power scaling logic320 may be one greater than the actual upper power bound so that theupper-bound compare result 336, when low, indicates that the proposedpower is above the upper power bound value 323 and, when high, indicatesthat the proposed power is below or equal to the upper bound power boundvalue 323.

Reflecting on the operation of the power scaling logic 320, it can beseen that the proposed power may, in some instances, be greater than theupper power bound or less than the lower power bound by more than one(e.g., if the power in the initial tap weights matches the upper powerbound and the magnitude of more than one tap weight is increased). Inone embodiment, this circumstance is tolerated, as iterative adjustmentof the primary tap weight will ultimately bring the applied power withinthe power constraint. In an alternative embodiment, the primary tapweight may be adjusted in each tap weight update cycle according todifference between the proposed power and upper power bound (or lowerbound), thereby ensuring that the power constraint will be met in eachupdate. In either embodiment, after the primary tap weight is adjusted,the complete set of updated tap weights may be provided to the transmitcircuit, for example, via the back channel 225 shown in FIG. 3.Alternatively, as with the residue-based power scaling logic of FIG. 8,the power scaling logic 320 may be implemented in the transmit-side ICdevice, with the update values (or error sign values and data signvalues) being provided via the back channel 225 or other signaling path.

Differential Samplers

FIG. 10 illustrates an embodiment of a differential sampler 360 that maybe used to implement the data sampler 211 and adaptive sampler 213 ofFIG. 3. The sampler 360 includes a preamplifier stage 361 and samplingstage 385. The preamplifier stage 361 includes a pair of differentialamplifiers 362 and 363 each biased by a respective pair of current DACs(IDACs) 380/382 and 384/386, and each having first and second outputnodes 378 and 379 coupled to a supply voltage via a respective resistiveelement, R. The resistive elements may be implemented, for example,using diode-configured transistors, biased transistors, resistors, orany other active or passive circuitry for establishing a resistance.Transistors 365 and 364 within differential amplifier 362 have widths W1and W2, respectively, with W1 being greater than W2. Transistors 368 and367 within differential amplifier 363 also have respective widths W1 andW2. A differential input signal composed of signal component signals x′and /x′ is provided to each of the differential amplifiers 362, 363 withx′ being provided to gate terminals of transistors 364 and 368 and /x′being provided to gate terminals of transistors 365 and 367. By thisarrangement, when control values C_(OFST) and C_(DLEV) (e.g., generatedby an adaptive module as described in reference to FIG. 7) aresubstantially equal to complement control values /C_(OFST) and/C_(DLEV), respectively (e.g., in an 8-bit control word,C_(DLEV)=C_(OFST)=128 and /C_(DLEV)=/C_(OFST)=127), the differentialamplifiers 362 and 363 are substantially balanced, operating in effectas a single differential amplifier having component transistors of widthW1+W2. Thus, if x′ is greater than /x′, transistors 364 and 368 willcollectively sink more current than transistors 365 and 367, therebycausing the voltage on output node 378 to be pulled down (i.e., via theresistive element, R, coupled to the output node 378) more than thevoltage on output node 379.

When the preamplifier stage 361 is balanced (i.e., control valuessubstantially equal to complement control values), the voltages on thepreamplifier output nodes 378 and 379 are substantially equal when inputsignals x′ and /x′ are at the common mode potential (i.e., as when x′and /x′ cross one another in transition). Thus, in the absence ofsystematic DC offset, the effective threshold of the preamplifier stage361, and therefore the sampler 360 as a whole, occurs at the common modeof x′ and /x′. By contrast, when the preamplifier is imbalanced, forexample, by increasing C_(DLEV) relative to /C_(DLEV), equal values ofx′ and /x′ result in output node 379 being pulled lower than output node378 due to the fact that transistor 365 is wider than transistor 364(and therefore has a greater gain), and that the compensating(balancing) effect of differential amplifier 363 is diminished by thereduced control value /C_(DLEV). Thus, increasing C_(DLEV) relative to/C_(DLEV) increases the effective threshold of the preamplifier abovethe common mode. By increasing C_(DLEV) to the point at which thethreshold between ‘0’ and ‘1’ signal levels is set to the target datalevel, DLEV, a sampler having a threshold level at DLEV is achieved. Byreversing the connections of the C_(DLEV) and /C_(DLEV) values to thecurrent DACs of a counterpart sampler (not shown), a sampler having athreshold level at −DLEV is achieved. Such a technique is applied in amulti-level signaling embodiment described below.

Still referring to the preamplifier stage 361, it should be noted thatin the case of a binary data sampler, such as element 211 of FIG. 3, thedesired threshold occurs at the common mode of the incoming data signals(i.e., the “zero” threshold). Accordingly, in a sampler dedicated tobinary data sampling, the current DACs 382 and 386 may be omitted orreplaced with fixed-bias, or self-biased current sources.

The sampling stage 385 includes a differential amplifier 397 formed bytransistors 398 and 399, a sense amplifier 387 formed by back-to-backcoupled inverters 388 and 389, and a storage circuit 396 formed by aset-reset flip-flop. The differential amplifier 397 includes controlinputs coupled to the output nodes 378 and 379, respectively, of thepreamplifier stage 361, and output nodes 391 and 393 coupled to sourceterminals of the inverters 388 and 389, respectively. A biasingtransistor 390, switchably controlled by the sampling clock signal 210(or other sample control signal), is coupled between the differentialamplifier 397 and a ground reference (or other low voltage reference).The sampling clock signal 210 is additionally coupled to control inputsof positively-doped MOS (PMOS) transistors 394 and 395 which are coupledbetween a supply voltage (e.g., V_(DD)) and output nodes of theinverters 388 and 389. By this arrangement, when the sampling clocksignal 210 is low, transistor 390 is switched off, and transistors 394and 435 are switched on to pre-charge the output nodes of the inverters388 and 389 to the supply voltage. The output nodes of the inverters 388and 389 are coupled to active-low set and reset inputs, respectively, ofthe storage circuit 396, so that the content of the storage circuit 396is maintained through the low half-cycle of the sampling clock signal210. When the sampling clock signal 210 goes high, biasing transistor390 is switched on and draws current through the two transistors 399 and398 of the differential amplifier 397 in proportion to the voltagesdeveloped on the output nodes 378 and 379 of the preamplifier stage 361.Thus, if the voltage developed on node 379 is higher than the voltage onnode 378, the current drawn by biasing transistor 390 will flowprimarily through transistor 398. Conversely, if the voltage developedon node 378 is higher than the voltage on 379, the current drawn bybiasing transistor 390 will flow primarily through transistor 398.Transistors 394 and 395 are switched off in response to the high-goingsampling clock signal 210 so that the pre-charged outputs of theinverters 388 and 389 are discharged by currents flowing throughtransistors 398 and 399. By this operation, if the incoming differentialsignal (x′) exceeds the common mode voltage, (i.e., (x′+/x′)÷2), by morethan the target data level threshold (i.e., the incoming differentialsignal exceeds the target threshold level, DLEV), the current drawn bybiasing transistor 390 will flow primarily through transistor 398.Consequently, the output node of inverter 389 will be discharged morerapidly than the output node of inverter 388, driving the output ofinverter 389 low and driving the output of inverter 388 high (i.e., thePMOS transistor within inverter 388 is switched on and the NMOStransistor within inverter 388 is switched off). The low output ofinverter 389 is applied to the active-low set input of the storagecircuit 396, causing the storage circuit 396 to store a logic ‘1’sampled data value. By contrast, if the incoming signal level does notexceed the target data level threshold, the current drawn by biasingtransistor 390 will flow primarily through transistor 399, therebydriving inverter 388 low (and driving inverter 389 high) to store alogic ‘0’ sampled data value within storage circuit 396.

Still referring to FIG. 10, during a DC offset calibration operation,null-valued differential signals are applied to the differential inputsof the preamplifier stage 361 either by transmission of null valued dataover the signaling path (i.e., x=/x), or by locally coupling thedifferential inputs to one another such that x′=/x′ (e.g., by activationof one or more pass-gate-configured transistors in response to acalibration signal). In the case of transmission of null valued data, ifa DC offset in the differential signals is induced by the signalingpath, or if the preamplifier stage 361 or sampler stage 385 havesystematic DC offsets (e.g., due to threshold voltage (V_(T)) mismatchesin the differential transistor pairs 364/365, 367/368 and/or 398/399),then the effective threshold of the sampler 360 will not occur at thecommon mode of x and /x (i.e., the transmit-side common mode).Similarly, in the case of local, switched coupling of differentialinputs (i.e., to force a common mode input to transistor pairs 364/365and 367/368), the effective threshold of the sampler 360 will not occurat the common mode if the preamplifier stage or sampler stage exhibitsystematic DC offsets. In either case, the non-common-mode threshold maybe detected in an offset calibration operation by the repeated positiveor negative sign of the sampled data, and the C_(OFST) value may beincremented or decremented (and /C_(OFST) correspondingly decremented orincremented) as discussed above to bias the sampler to a calibratedstate.

FIG. 11 illustrates an embodiment of a current DAC 381 that may be usedto implement the current DACs 380, 382, 384 and/or 386 within thesampler 360 of FIG. 10, and/or the current DAC 237 within the outputdriver of FIG. 5. The current DAC 381 includes control transistors 407₀–407 _(N−1) and biasing transistors 409 ₀–409 _(N−1). Each of thecontrol transistors 407 ₀–407 _(N−1) is coupled in series (e.g., sourceto drain) with a corresponding one of the biasing transistors 409 ₀–409_(N−1) to form a transistor pair that is coupled between a referencevoltage (ground in this example) and an output node 408 (i.e., the nodeto be connected to the source terminals of the transistors which formthe differential amplifier 362 of FIG. 10). Gate terminals of thecontrol transistors 407 ₀–407 _(N−1) are coupled to receive respectivecomponent signals, C[0]–C[N−1], of a multi-bit control value, such as adata level threshold, DC offset setting, tap weight, or other controlvalue. Each of the control transistors 407 ₀–407 _(N−1) has a binaryweighted gain such that a current of I_(REF)×2^(i) (where i representsthe i^(th) transistor in the positions 0, 1, 2, . . , N−1) flows throughcontrol transistor 407 _(i) when the corresponding control signalcomponent is high. Thus, if all the constituent bits of the controlvalue C[N−1:0] are high, then I_(REF) flows through control transistor407 ₀, I_(REF)×2 flows through transistor 407 ₁, I_(REF)×4 flows throughcontrol transistor 407 ₂, and so forth to control transistor 407 _(N−1)which conducts I_(REF)×2^(N−1). Accordingly, control transistors 407₀–407 _(N−1) are designated ×1, ×2 . . . , ×2^(N−1) transistors,respectively. By this arrangement, the control value C[N−1:0] may be setto any of 2^(N) values to select bias currents that range from 0 toI_(REF)×2^(N−1) in increments of I_(REF). The biasing transistors 409₀–409 _(N−1) have gate terminals coupled to receive a bias voltage,V_(BIAS), that is adjusted as necessary (e.g., by a biasing circuit) toestablish or maintain a desired I_(REF).

In one embodiment, the relative gains (i.e., transconductance values) ofthe various transistors used to implement the current DAC 381 areestablished by adjusting the width-length ratio (i.e., W/L) ofindividual control transistors 407 and/or biasing transistors 409. Forexample, the width-length ratio of the ×2 control transistor 407, istwice the width-length ratio of the ×1 control transistor 407 ₀, thewidth-length ratio of the ×4 control transistor 407 ₂ is twice thewidth-length ratio of the ×2 control transistor 407 ₁, and so forth. Thebiasing transistors 409 may have similar gain ratios relative to oneanother (e.g., ×1, ×2, ×4, ×2^(N−1) as shown in FIG. 11). Othertechniques for adjusting the relative gains of the control transistors407 and biasing transistors 409 may be used in alternative embodiments.Also, weightings other than binary weightings may be used. For example,in one embodiment, each of the control transistors 407 has an equal gainto each of the other control transistors 407 such that the current drawnby the current DAC 381 is proportional to the number of logic ‘1’ bitsin the control value, C[N−1:0].

FIG. 12 illustrates an alternative embodiment of a sampler 420 that maybe used to implement the data sampler 211 and adaptive sampler 213 ofFIG. 3. The sampler 420 includes a sampling stage 422 and an offsetcontrol circuit 410. The sampling stage 422 is implemented in generallythe same manner as the sampling stage 385 of FIG. 10 (and includesdifferential amplifier 397, sense amplifier 387, biasing transistor 390,and storage circuit 396), except that the input signal lines carrying x′and /x′ are coupled directly to the control terminals of transistors 398and 399, respectively. The offset control circuit 410 includes adifferential amplifier 418 having output nodes coupled to nodes 391 and393 of the sampling stage 422. Control terminals of transistors 417 and419 of the differential amplifier 418 are biased by respective voltageDACs 425 and 427. Voltage DAC 427 includes current DACs 415 and 416coupled to a resistive pull-up element 423 and controlled by controlvalues C_(DLEV) and C_(OFST), respectively. Voltage DAC 425 similarlyincludes current DACs 413 and 414 coupled to a resistive pull-up element421 and controlled by complement control values /C_(DLEV) and /C_(OFST).By this arrangement, when the sampling clock signal 210 goes high, thecurrent through output node 393 of the sampling stage 422 is a sum ofthe currents drawn by transistor 398 of the sampling stage 422 andtransistor 417 of the offset control circuit 410. Similarly, the currentthrough node 391 of the sampling stage 422 is a sum of the currentsdrawn by transistor 399 of the sampling stage 422 and transistor 419 ofthe offset control circuit 410. As discussed above in reference to FIG.17, when the current through node 391 exceeds the current through node393, a logic ‘1’ is stored within storage circuit 396 and, conversely,when the current through node 393 exceeds the current through node 391,a logic ‘0’ is stored within storage circuit 396.

When the complementary DAC control values C_(DLEV) and /C_(DLEV), andC_(OFST) and /C_(OFST) are substantially the same, and in the absence ofDC offset, the sampler 420 is balanced and the effective thresholdoccurs at the common mode of the incoming x′ and /x′ signal levels. Thatis, if x′ exceeds the common mode voltage, V_(CM)=(x′+/x′)÷2, thecurrent through node 393 exceeds the current through node 391, causing alogic ‘1’ to be captured as the sampled data value. As C_(DLEV) isincreased and /C_(DLEV) correspondingly decreased, the effectivethreshold of the differential amplifier is increased such that x′ mustbe higher than /x′ by an amount necessary to overcome the additionalcurrent drawn by transistor 419 of the offset control circuit 410. Thus,by increasing C_(DLEV) and decreasing /C_(DLEV), the effective thresholdof the sampling circuit 420 may be set to the target data levelthreshold. That is, a logic ‘1’ is output as the sampled data value ifthe difference between x′ and /x′ exceeds the target data levelthreshold, and a logic ‘0’ is output otherwise. By reversing theconnections of the C_(DLEV) and /C_(DLEV) values to the current DACs 416and 414, a sampler having a threshold level at −DLEV is achieved. Such atechnique is applied in a multi-level signaling embodiment describedbelow.

As with the sampler 360 of FIG. 10, during an offset calibrationoperation within the sampler 420, null-valued differential signals areapplied to the differential inputs of the sampling stage 422 either bytransmission of null valued data over the signaling path (i.e., x=/x),or by locally coupling the differential inputs to one another such thatx′=/x′ (e.g., by activation of one or more pass-gate-configuredtransistors in response to a calibration signal to switchably couple thegates of transistors 398 and 399). In either case, the non-common-modethreshold may be detected in an offset calibration operation by therepeated positive or negative sign of the sampled data, and the C_(OFST)value may be incremented or decremented (and /C_(OFST) correspondinglydecremented or incremented, respectively) to bias the sampler 420 to acalibrated state.

Still referring to FIG. 12, in the case of a binary data sampler such assampler 211 of FIG. 3, the desired threshold occurs at the common modeof the incoming data signals (i.e., the “zero” threshold). Accordingly,in a sampler dedicated to binary data sampling, the current DACs 414 and416 may be omitted or replaced with fixed-bias, or self-biased currentsources.

Updating Tap Weights in Response to Data Level Error

FIG. 13 is a canonical diagram of a channel 431 and receive-sideequalizer 433 that may be used to adaptively determine a set ofequalizer tap weights. An input signal, x(n), is transformed as itpropagates through the channel, yielding a channel response, u(n) which,in turn, is operated upon by the receive-side equalizer 433 to produce asystem response, x′(n). The system response is input to a sampler 435(or comparator) which subtracts a delayed version of the originallytransmitted signal (−x(n−dly)) from the system response to produce anegative error signal, −e(n). Thus, the error signal e(n) represents thedifference between the originally transmitted signal, x(n) and systemresponse x′(n) and is negative when system response exceeds theoriginally transmitted signal and positive when the originallytransmitted signal exceeds the system response. Together, the channelresponse and the error signal may be used to update the equalizer tapweights, for example, through application in a least mean square errordetermination.

Assuming a linear channel response, the linear filtering effect of theequalizer is commutative and therefore may be applied to the inputsignal, x(n), before the signal is transmitted on the channel 431. Thatis, instead of receive-side equalization, transmit-side pre-emphasis maybe used to establish a pre-emphasized input signal, y(n) which, afterpropagating through the channel 431, yields a system response x′(n) thatcorresponds to the system response x′(n) realized in the receive-sideequalization system of FIG. 13. Unlike the receive-side equalizingsystem of FIG. 13, however, the channel response is generallyunavailable to the transmit side of the signaling system, complicatingtap weight update operations. In one embodiment of the invention,depicted in the canonical diagrams of FIGS. 14A and 14B, a two phaseapproach is used to update the tap weights. In the first phase, transmitpre-emphasis taps within a transmit circuit (i.e., post-taps andpre-taps) are disabled so that the input signal, x(n) is unmodified bythe transmit pre-emphasis circuitry 441, and propagates through thechannel 431 to produce a channel response u(n). By this operation, thechannel response, u(n), is effectively pre-computed by the channel 431itself. In the embodiment of FIG. 14A, the channel response, u(n), isreturned to the transmit-side device (e.g., through a back channel orother communication path), where it is stored for later application in atap weight update operation. Alternatively, the channel response, u(n),is stored by the receive-side device. After the channel response hasbeen obtained, the second phase of the tap weight operation is begun byenabling the pre-emphasis circuitry 441, and then re-transmitting theinitial signal, x(n). In the second phase, the pre-emphasis circuitry441 modifies the initial signal, x(n), to generate a pre-emphasizedsignal, y(n), which, in turn, propagates through the channel 431 togenerate the system response, x′(n). The system response, x′(n), iscompared with the delayed version of the initial signal (the delaycorresponding, for example, to channel propagation time) to generate anerror signal, −e(n). In the embodiment of FIG. 14B, the error signal isprovided to the transmit-side device where it is applied, along with thepreviously stored channel response, u(n), in a tap weight updateoperation. Alternatively, if the channel response is stored in thereceive-side device, the error signal and channel response may beapplied by the receive side device to generate a set of tap weightupdate values, or a set of updated tap weights. The update values (ortap weights) are then returned to the transmit side device and used toupdate the existing tap weights applied within the pre-emphasiscircuitry 441 (or, in the case of updated tap weights, substituted forthe existing tap weights).

In one embodiment, the tap weight update operation is a sign-sign LMSoperation in which the sign of the channel response and sign of theerror signal are used to update the tap weights as follows:W _(N+1) =W _(N)+stepsize*sign(e _(n))*sign( u _(n))   (9).Thus, only the signs of the channel response and error signal need bereturned to the transmit-side device (or stored in the receive sidedevice) in the first and second phases of a tap weight update operation.After the transmit pre-emphasis tap weights have been updated, two-phasetap weight update operations are repeated as necessary for thepre-emphasis tap weights to converge to a setting that corresponds to aminimum (or near-minimum) mean square error, and thereafter tocompensate for system drift (e.g., due to changes in voltage andtemperature). Note that by updating the tap weights in this way, thereceiver response is included in the channel response.

FIG. 15 is a flow diagram of the two-phase tap weight update operationdescribed in reference to FIGS. 14A and 14B. Initially, at start block449, an index, n, that indicates the number of completed tap weightupdates is initialized to zero. At block 451, the transmit pre-emphasiscircuitry is disabled. At block 453, a first sequence of data values,referred to herein as training sequence(n), is transmitted over thechannel (e.g., a differential or single-ended signal path) to generatethe channel response u(n). At block 455, the transmit pre-emphasiscircuitry is enabled so that an initial setting of tap weights (i.e., inthe first iteration) are applied to generate the pre-emphasized signaly(n) illustrated in FIG. 14B. In one embodiment, the initial setting oftap weights includes zero-valued pre- and post-tap weights, and amaximum-valued primary tap weight. In alternative embodiments, theinitial setting of tap weights may be determined according to systemcharacteristics or empirical determination of a desired tap weightsetting. At block 457, training sequence(n) is re-transmitted togenerate a system response, x′(n) and corresponding error signal, e(n).At block 459, tap weight updates (i.e.,stepsize*sign(u_(n))*sign(e_(n))), or updated tap weights themselves(i.e., W _(n+1)) are generated based on the channel response and errorsignal. At block 461, the tap weight updates generated in block 459 areapplied to update the existing tap weights (or the updated tap weightsgenerated in block 459 are substituted for the existing tap weights),and, at block 463, the index variable, n, is incremented to indicatethat a first tap weight update has been completed.

In the embodiment of FIG. 15, an overall tap weight adaptation operationinvolves iteratively performing the operations of blocks 451–463 apredetermined number of times. In such an embodiment, the indexvariable, n, is evaluated at decision block 465 to determine if n hasbeen incremented past a predetermined value. If so, the tap weightupdate operation is deemed to be complete. In an alternative embodiment,the operations of blocks 451–463 are repeated until tap weight updatesresult in negligible reduction in the error signal. In anotheralternative embodiment, the operations of blocks 451–463 are repeateduntil all or a subset of the tap weights are determined to be ditheringby one or more steps.

Reflecting on the adaptive generation of pre-emphasis tap weightsachieved by iteratively performing the two-phase tap weight updatesdescribed in reference to FIGS. 14A, 14B and 15, it can be seen that therepeated determination of the channel response, u(n), enables astatistical approximation of random noise. That is, in the absence ofrandom noise, like channel responses will be obtained in block 453 forlike training sequence transmissions. Thus, by iteratively performingthe two-phase tap weight updates described in reference to FIGS. 14A,14B and 15, the pre-emphasis tap weights effectively converge tosolution that represents a minimum (or near minimum) mean squared error.

FIG. 16 illustrates a single-phase tap weight adaptation approach thatneglects the effects of noise, and therefore constitutes a zero-forcingsolution. Rather than disabling the pre-emphasis circuitry 441 as in thefirst phase of the two-phase operation of FIGS. 14A and 14B, thepre-emphasis circuitry 441 is left enabled to generate a pre-emphasizedinput signal, y(n) which, after propagating through the channel 431,yields a system response x′(n) that corresponds to the system responserealized in the receive-side equalization system of FIG. 13. The systemresponse is compared with a delayed version of the input signal (i.e.,−x(n−dly)) to generate an error signal, −e(n). The system response anderror signal are then supplied to the transmit side device and appliedin a tap weight update operation. As in the two-phase approach, the tapweight update operation may alternatively be performed in thereceive-side device and tap weight updates, or updated tap weightsthemselves communicated to the transmit-side device (e.g., via a backchannel). In one embodiment, the signs of the system response and errorsignal are applied in the tap weight update operation in accordance withexpression (3) above (i.e., a sign-sign LMS update operation). By thisoperation the pre-emphasis tap weights are iteratively adjusted toachieve a zero-forcing solution.

FIG. 17 is a flow diagram of the single-phase, zero-forcing tap weightupdate operation described in reference to FIG. 16. Initially, at startblock 471, an index, n, that indicates the number of completed tapweight updates is initialized to zero. At block 473, the transmitpre-emphasis circuitry 441 of FIG. 16 is enabled, for example, byestablishing an initial set of tap weights. In one embodiment, theinitial setting of tap weights includes zero-valued pre- and post-tapweights, and a maximum-valued primary tap weight. In alternativeembodiments, the initial setting of tap weights may be determinedaccording to system characteristics or empirical determination of adesired tap weight setting. After the transmit pre-emphasis circuitry isenabled, a first training sequence(n), is input to the transmitpre-emphasis circuit at block 475 to establish a pre-emphasized inputsignal, y(n), which, after propagating through the channel, yields asystem response x′(n) and, upon comparison of x′(n) with x(n−dly), anerror signal e(n). At block 477, tap weight updates (i.e.,stepsize*sign(x′n)*sign(e_(n))), or updated tap weights themselves(i.e., W _(n+1)) are generated based on the system response and errorsignal. At block 479, the tap weight updates generated in block 477 areapplied to update the existing tap weights (or the updated tap weightsgenerated in block 477 are substituted for the existing tap weights),and, at block 481, the index variable, n, is incremented to indicatethat a first tap weight update has been completed.

In one embodiment, the operations of blocks 475–481 are repeated until,at decision block, 483, the index variable, n, is determined to havereached a final value. When the final value is reached, the tap weightadaptation operation is deemed completed. In an alternative embodiment,the operations of blocks 475–481 are repeated until tap weight updatesresult in negligible reduction in the error signal. In anotheralternative embodiment, the operations of blocks 475–481 are repeateduntil all or a subset of the tap weights are determined to be ditheringby one or more steps.

Referring again to FIG. 16, by using an adapted, target threshold levelto generate error signals, rather than x(n−dly), live data rather thanpre-selected training sequences, may be used to adapt the tap weights.In one embodiment, for example, the adaptive sampler 213 of FIG. 3 isused to generate the error signal used to update the tap weights, withthe error signal being filtered according to whether the correspondingdata sign value (i.e., sign of x′_(n)) indicates a system responsehaving a state that should match the data level. As another example, theerror signal may be filtered according to desired partial-response datasequences (e.g., searching for bit sequences ‘11’, ‘00’, ‘111’, ‘000’,or longer sequences depending on the number of ISI components in thepartial response).

Adaptive Sampler as Proxy Data Sampler

FIG. 18 illustrates a multi-sample receiver 500 according to anembodiment of the invention. The receiver 500 includes a data sampler501, adaptive sampler 503 and adaptive module 505 that are implementedin substantially the same manner as the samplers 211, 213 and adaptivemodule 215 of FIG. 3, except that the adaptive module 505 includescircuitry for generating offset cancellation values, OFST_(D) andOFST_(A), for the data sampler 501 and adaptive sampler 503,respectively (e.g., as described in reference to FIG. 7). The receiver500 additionally includes a pair of threshold multiplexers 507 and 509,and a pair of output path multiplexers 511 and 513. The thresholdmultiplexers 507 and 509 enable the threshold values supplied to thedata sampler and adaptive sampler to be swapped such that the datasampler receives the data level threshold, DLEV, generated by theadaptive module 505, and the adaptive sampler receives a zero threshold.Similarly, the output path multiplexers 511 and 513 enable the adaptivemodule inputs driven by the data sampler 501 and adaptive sampler 503 tobe swapped such that the adaptive sampler 503 provides a sample value tothe data sign input of the adaptive module (and therefore drives thereceive data path), and the data sampler 501 provides a sample value tothe error sign input of the adaptive module. By this arrangement, thefunctions of the adaptive sampler 503 and data sampler 501 may beswapped. In particular, the adaptive sampler 503 may act as a proxy forthe data sampler 501, enabling continued reception of data, while thedata sampler 501 is taken out of service for testing, calibration or anyother activity that would ordinarily interrupt data reception.

In the embodiment of FIG. 18, a mode select signal, referred to hereinas a proxy-enable signal 516 (PE), is used to select between normal andproxy modes of operation within the receiver 500 and is coupled to thecontrol inputs (i.e., select inputs) of the threshold multiplexers 507and 509, and the output path multiplexers 511 and 513. Each of themultiplexers 507, 509, 511 and 513 has first and second input ports(i.e., designated ‘0’ and ‘1’, respectively, in FIG. 18), with thesignal present at the first input port being selected and output fromthe multiplexer in response to a logic low proxy-enable signal 516 andthe signal present at the second input port being selected and outputfrom the multiplexer in response to a logic high proxy-enable signal516. A zero threshold is supplied to the first input port of thresholdmultiplexer 507 and to the second input port of threshold multiplexer509, and the target data level threshold, DLEV, generated by theadaptive module 505 is supplied to the second input of thresholdmultiplexer 507 and to the first input port of threshold multiplexer509. By this arrangement, when the proxy-enable signal 516 is low,enabling the normal operating mode of the receiver 500, the zerothreshold is output from threshold multiplexer 507 and the data levelthreshold is output from threshold multiplexer 509. Conversely, when theproxy-enable signal 516 is high, enabling the proxy mode of operationwithin receiver 500, the zero threshold is output from thresholdmultiplexer 509 and the data level threshold is output from thresholdmultiplexer 507. In one embodiment, the thresholds output from thethreshold multiplexers 507 and 509 are summed with the offsetcancellation values OFST_(A) and OFST_(D) in summing circuits 515 and517, respectively (e.g., digitally summed, or current sum) to generatethe thresholds supplied to the data and adaptive samplers 501 and 503.Thus, in the normal mode, the data sampler generates a data sign value216, sgn(x′_(n)) that indicates whether the incoming signal, x′_(n), isgreater or less than the zero threshold (e.g., offset-calibrated commonmode), and the adaptive sampler 503 generates an error sign value 218,sgn(e_(n)), that indicates whether the incoming signal, x′_(n) isgreater or less than the target data level threshold, DLEV. That is, inthe normal mode, the data and adaptive samplers 501 and 503 generatedata sign and error sign values in the manner described in reference toFIG. 3. By contrast, in the proxy mode, the roles of the data andadaptive samplers 501 and 503 are reversed, with the adaptive sampler503 operating as a proxy for the data sampler 501 to generate a datasign value and vice-versa.

The output path multiplexers 511 and 513 each have first and secondinput ports coupled to receive the outputs of the data sampler 501 andadaptive sampler 503. More specifically, the first input port of outputpath multiplexer 511 and the second input port of output pathmultiplexer 513 are coupled to the output of the data sampler 501, andthe second input port of output path multiplexer 511 and the first inputport of output multiplexer 513 are coupled to the output of the adaptivesampler 503. By this arrangement, when the receiver 500 is in the normalmode, the data sign values 216 generated by the data sampler 501 areprovided to the data sign input of the adaptive module 505, and theerror sign values 218 generated by the adaptive sampler 503 are providedto the error sign input of the adaptive module 505. Conversely, in theproxy mode, the data sign values generated by the adaptive sampler 503are provided to the data sign input of the adaptive module 505 and theerror sign values generated by the data sampler 501 are provided to theerror sign input of the adaptive module 505.

In many applications, once the data level threshold, DLEV, has convergedto the target level, the data level threshold changes relatively slowly,for example, in response to voltage and temperature drift. Consequently,the stream of error sign values delivered to the adaptive module 505 maybe temporarily interrupted without significant adverse impact on thereceiver 500 or the signaling system as a whole. By contrast, if thestream of data sign values is interrupted, the communication link (e.g.,over signaling path 202) is lost for the duration of the interruption.By placing the receiver 500 in proxy mode, and thereby swapping theroles of the data and adaptive samplers 501 and 503, the data sampler501 may be temporarily removed from service without interrupting datareception. In one embodiment, for example, an offset calibrationoperation is performed by switching the receiver 500 to proxy mode(i.e., asserting the proxy enable signal 516); temporarily zeroing thedata level threshold, DLEV; switchably coupling the differential inputsof the data sampler 501 to one another (and switchably isolating theinputs from the signal path 202 so as not to short the component signallines of the signal path to one another); then adjusting the OFST_(D)value until the sample value generated by the data sampler 501 begins todither between ‘1’ and ‘0’ states. The dithering sample value indicatesthat the null signal input to the data sampler 501 is being detected andtherefore that the offset calibration is complete. After completing theoffset calibration for the data sampler 501, the control setting for thedata level threshold is restored, and the proxy-enable signal 516 islowered to re-establish the normal operating mode of the receiver 500.At this point, the data sampler 501 has been removed from service forcalibration purposes, then restored to service without interruption indata reception.

Still referring to FIG. 18, the proxy mode of the receiver 500 may alsobe used to more permanently swap the roles of the data and adaptivesamplers 501 and 503, in effect establishing the adaptive sampler 503 asthe full time data sampler, and the data sampler 501 as the full-timeadaptive sampler. This may be desirable, for example, if it isdetermined that the adaptive sampler exhibits a lower bit error rate,less jittery output, lower DC offset, or other characteristicimprovement relative to the data sampler 501.

A number of changes may be made to the embodiment of FIG. 18 withoutdeparting from the scope of the present invention. For example, if theproxy mode is to be used only to enable the adaptive sampler 503 tostand-in for the data sampler 501, then the threshold multiplexer 507may be omitted. Offset calibration is simplified in such anarchitecture, as the data level threshold is not supplied to the datasampler 501 in proxy mode and therefore need not be zeroed. In analternative embodiment, the threshold multiplexers 507 and 509 may becontrolled by separate signals so that, if an offset calibration is tobe performed in the data sampler 501, only the threshold input to theadaptive sampler 503 is switched (i.e., by selecting the zero thresholdto be supplied to the adaptive sampler 503), so that the data sampler501 continues to receive the zero threshold, obviating the temporaryzeroing of the data level threshold. Such an embodiment has theadditional benefit of enabling both the data sampler 501 and theadaptive sampler 503 to generate sign data values simultaneously, forexample, for confirmation of accurate data reception (a third samplermay be provided for voting purposes). Separate control signals may alsobe provided to the output path multiplexers 511 and 513 so that the dataand adaptive samplers 501 and 503 can be enabled to simultaneouslygenerate data sign values for a given time period before switching theoutput path multiplexer 511 to select the adaptive sampler 503 toprovide data sign values to the adaptive module 505. In this manner, amake-before-break operation is enabled within the receive circuit 500,instead of abruptly transitioning between the adaptive and data samplers501 and 503 as the source of data sign values.

Tap Weight and Data Level Adaptation in a Multi-Level Signaling System

FIG. 19 illustrates a multi-level signaling system 530 according to anembodiment of the invention. The multi-level signaling system 530includes a multi-level, multi-tap transmitter 531, and a multi-level,multi-sample receiver 539, coupled to one another via high-speedsignaling path 532. As in the signaling system of FIG. 3, the signalpath 532 may be a differential signaling path having a pair of componentsignal lines to conduct differential multi-level signals generated bythe transmitter 531, or a single-ended signaling path for transmissionof single-ended multi-level signals generated by the transmitter 531.Also, the signal path 532 may be formed in multiple segments disposed ondifferent layers of a circuit board and/or multiple circuit boards(e.g., extending between backplane-mounted daughterboards, betweenmotherboard and daughterboard, etc.). In one embodiment, the transmitter531 and receiver 539 are implemented in respective integrated circuit(IC) devices that are mounted on a common circuit board or differentcircuit boards (e.g., as in the case of backplane-mounteddaughterboards). In alternative embodiments, IC dice (i.e., chips)containing the transmitter 531 and receiver 539 may be packaged within asingle, multi-chip module with the chip-to-chip signaling path formed bybond wires or other signal conducting structures. Also, the transmitter531 and receiver 539 may be formed on the same IC die (e.g., system onchip) and the signaling path 532 implemented by a metal layer or otherconducting structure of the die.

In the embodiment of FIG. 19, the transmitter 531 includes a transmitshift register 533, output driver bank 534 and tap weight register 536,and generates output signals having one of four pulse amplitudemodulation levels (i.e., 4-PAM) according to the state of a two-bittransmit data value (received, for example, by a two-line inputdesignated “TX DATA”). In the particular embodiment shown, the transmitshift register 533 is five elements deep and used to store a pre-tapdata value D₊₁, primary data value D₀, and three post-tap data valuesD⁻¹, D⁻² and D⁻³, with each of the pre-tap, post-tap and primary-datavalues having two constituent bits. As in the transmit circuit 201 ofFIG. 3, the primary data value is the data value to be transmitted(i.e., communicated) to the receiver 539 during a given transmissioninterval, and the pre- and post-tap data values are the next-to-betransmitted and previously transmitted data values, respectively (i.e.,the subscript indicating the number of transmission intervals totranspire before the data value will be transmitted). Each of the shiftregister storage elements is coupled to a respective one of multi-leveloutput drivers 535 ₀–535 ₄ within the output driver bank 534, withoutput driver 535 ₁ forming the primary driver, output driver 535 ₀forming the pre-tap driver and output drivers 535 ₂–535 ₄ forming thepost-tap drivers. Different numbers of pre- and post-tap drivers may beused in alternative embodiments.

As in the transmit circuit of FIG. 3, the tap weight register 536 isused to store the tap weights W_(N)(0)–W_(N)(4) supplied to the outputdrivers 535 ₀–535 ₄, respectively, with updated tap weights W_(N+1) 236being supplied by the multi-level receiver 539, for example, via a backchannel 225. In one embodiment, the signal path 532 is pulled up to apredetermined voltage level (e.g., at or near supply voltage) bysingle-ended or double-ended termination elements, and the outputdrivers 535 ₀–535 ₄ generate multi-level signals (i.e., symbols) on thesignal path 532 by drawing a pull-down current, I_(PD) (i.e., dischargecurrent), in accordance with the corresponding tap weight and datavalue. More specifically, in one embodiment, the pull-down currentgenerated by the output driver corresponds to the most- andleast-significant bits (MSB and LSB) of a two-bit data value, D₀, asfollows (I_(NOM) being a nominal full-scale current):

TABLE 2 Normalized D₀[1] D₀[0] Signal (MSB) (LSB) I_(PD) Level 0 0 0 +10 1 I_(NOM)/3 +1/3 1 1 2I_(NOM)/3 −1/3 1 0 I_(NOM) −1As in the embodiment of FIG. 3, the primary driver 535 ₁ is used totransmit, D₀, the two-bit data value to be transmitted during a givensymbol time, and the pre-tap and post-tap drivers are used to providetransmit pre-emphasis as necessary to reduce dispersion-type ISI andother low-latency distortion effects.

FIG. 20 illustrates an embodiment of a multi-level output driver 570that operates in accordance with Table 2, and which may be used toimplement each of the multi-level output drivers 535 ₀–535 ₄ of FIG. 19.The output driver 570 includes a pair of logic gates 571 and 573 andthree components 575, 577 and 579, and receives the MSB and LSB of atwo-bit data value, D[1:0], and tap weight, W_(N)(i), as inputs. Thelogic gates 571 and 573 convert the MSB and LSB inputs into componentdriver input signals, A, B and C according to the following logic table:

TABLE 3 MSB LSB A B C 0 0 0 0 0 0 1 1 0 0 1 1 1 1 0 1 0 1 1 1That is, A is asserted (i.e., to a logic ‘1’) if either the MSB or LSBis a logic ‘1’ (i.e., A=MSB+LSB, the ‘+’ symbol indicating a logicalOR), B is asserted if the MSB is a logic ‘1’ (i.e., B=MSB is a logic ‘1’and the LSB is a logic ‘0’ (i.e., C=MSB/LSB). The component driver inputsignals, A, B and C, are input to the component drivers 575, 577 and579, respectively, and the tap weight, W_(N)(i), is input to each of thecomponent output drivers.

In one embodiment, each of the component output drivers 575, 577 and 579is implemented by the circuit illustrated in FIG. 5 (other output drivercircuits may be used in alternative embodiments). A single pair ofresistive elements may be provided and shared between the componentoutput drivers 575, 577 and 579 (i.e., instead of three sets of theresistive elements designated ‘R’ in FIG. 5), or, as discussed inreference to FIG. 5, the resistive elements may be implemented bytermination elements coupled to the component lines of the differentialsignaling path. Each of the component output drivers 575, 577 and 579may additionally be biased (e.g., by a biasing circuit not shown) todraw substantially the same current, I_(NOM)/3, from the signaling path.By this arrangement, the currents drawn by the component output drivers575, 577 and 579 are cumulative so that the four different currentlevels illustrated in table 2 are generated for the corresponding statesof the MSB and LSB. That is, the four possible states of a two-bittransmit value are signaled on the signaling path by drawing I_(NOM)/3in none, one, two or three of the component drivers 575, 577 and 579, asillustrated in the following table:

TABLE 4 MSB LSB A B C I_(PD) 0 0 0 0 0 0 0 1 1 0 0 I_(NOM)/3 1 1 1 1 02I_(NOM)/3 1 0 1 1 1 I_(NOM)

Referring again to FIG. 19, the multi-level, multi-sample receivecircuit 539 includes a multi-level sampler 541, and an adaptive sampler543. The multi-level sampler 541 itself includes component samplers 561,563 and 565, that operate in generally the same manner as the data andadaptive samplers described above (e.g., in reference to FIGS. 3 and10–12) to output a sample value having a sign according to whether theinput signal, x′_(n) is greater or less than a threshold level. Two ofthe component samplers 561 and 565 are used to resolve the LSB of theincoming 4-PAM signal, and have thresholds set at counterpart thresholdlevels, T+ and T−, above and below a zero threshold. Component samplers561 and 565 are referred to herein as the positive LSB sampler (L+) andnegative LSB sampler (L−), respectively. The remaining component sampler563, referred to herein as the MSB sampler, receives (or is set to) thezero threshold and is used to resolve the MSB of the incoming 4-PAMsignal.

Referring to FIG. 21, the zero threshold is nominally set midway betweenthe normalized +/−1 signal levels that correspond to data states ‘00’and ‘11’, and midway between the corresponding +⅓ and −⅓ signal levelsthat correspond to data states ‘01’ and ‘11’. Thus, if the output of theMSB sampler is high, the MSB of the recovered data value is high. Thethreshold supplied to the positive LSB sampler 561 (i.e., T+) is setmidway between the normalized +1 and +⅓ signal levels (i.e., at thenormalized +⅔ level), and the threshold supplied to the negative LSBsampler 565 (i.e., T−) is set midway between the normalized −1 and −⅓signal levels (i.e., at the normalized −⅔ level). Consequently, if theLSB of a transmitted data value is a ‘0’ (i.e., a ‘10’ or a ‘00’ istransmitted), then the sample values generated by positive and negativeLSB samplers 561 and 565 will have the same state, either high or low,as the incoming signal level will either exceed both the T+ and T−thresholds (D=‘10’) or fall below both the T+ and T− thresholds(D=‘00’). By contrast, if the LSB of the transmitted data value is a‘1’, then the sample values generated by positive and negative LSBsamplers 561 and 565 will have different states, as the incoming signalwill exceed the T− threshold, but not the T+ threshold. Thus, the LSB ofthe recovered data value may be generated by exclusive-ORing the L+ andL− outputs.

Returning to FIG. 19, exclusive-OR gate 567 is coupled to receive theoutputs of the positive and negative LSB samplers 561 and 565 andgenerates the LSB sample for that incoming data signal. Thus, duringeach signal reception interval, the multi-level sampler 541 generates anMSB/LSB sample pair which is provided to the adaptive module 545. Theadaptive module 545 generates an error value 538 that indicates whetherthe incoming signal x′_(n) exceeds a threshold value, TA. In oneembodiment, the threshold value corresponds to the normalized +⅓ signallevel, thereby enabling generation of a DAC control value which may beleft shifted by one bit (i.e., multiplied by two) to generate the T+threshold (i.e., +⅔), and then complemented to generated the T−threshold (i.e., −⅔). In an alternative embodiment, discussed below, theT+ threshold may be generated by determining and then averaging thenormalized +1 and +⅓ signal levels. In another embodiment, discussedbelow, the T+ threshold may be determined directly, by sampling theincoming signal at the midpoint of transitions between +1 and +⅓ levels.In yet other embodiments, the normalized received signal levels may bedifferent than ±⅓ and ±1, such that the desired threshold levels (T+,T−) may be different than ±⅔ (e.g., being set at the midpoint betweenadjacent signal levels or at other points that improve signalingmargins, bit error rate or other system performance metric). In thisregard, the references to normalized signal levels herein are butexamples. Other signal levels and threshold levels may be used. In allsuch embodiments, the counterpart threshold, T−, may be generated bycomplementing (or inverting) the T+ threshold. Alternatively, the T−threshold may be independently generated by determining andleft-shifting the −⅓ threshold, by determining and averaging the −⅓ and−1 thresholds, or by sampling the incoming signal at the midpoint oftransitions between −1 and −⅓ levels.

Still referring to FIG. 19, the adaptive module 545 generates thethresholds, T+ and T−, provided to the multi-level sampler 541, thethreshold, TA, provided to the adaptive sampler 543, and respectiveoffset cancellation values, OFST_(A), OFST_(L'), OFST_(M) and OFST_(L−),for the adaptive sampler 543 and each of the component samplers 561, 563and 565 of the multi-level sampler 541. In alternative embodiments, allor a portion of the offset cancellation circuitry within the adaptivemodule 545 may be omitted so that offset cancellation values are notgenerated for the adaptive sampler 543 and/or component samplers 561,563 and 565. Also, one or more of the offset cancellation values,OFST_(A), OFST_(L+), OFST_(M) and OFST_(L−), may be shared between anytwo or more of the samplers 543, 561, 563 and 565.

FIG. 22 illustrates an embodiment of an adaptive module 600 that may beused to implement the adaptive module 545 shown in FIG. 19. The adaptivemodule 600 includes an MSB register 601, LSB register 605, error signregister 603, sign multiplier 609, finite state machine 607, powerscaling logic 611, filter 625, threshold counter 613, thresholdmultiplier 617, threshold inverter 619, offset counter 615, offsetregisters 627, 629, 631 and 633, error signal multiplexer 621 anddemultiplexer 623. The adaptive module 600 operates similarly to theadaptive module 250 of FIG. 7, with data sign values, MSB_(n), and errorsign values, e_(n), being loaded into the MSB register 601 and errorsign register 603, respectively, in response to a sampling clock signalor other control signal. LSB values, LSB_(n), are similarly loaded intothe LSB register 605. In the embodiment of FIG. 22, the MSB register 601is a five-deep shift register to store the most recently generated datasign values, MSB_(n−1)–MSB_(n−5), (other depths may be used inalternative embodiments) and outputs the data sign values to the signmultiplier 609. The sign multiplier 609 receives the data sign valuesfrom the MSB register 601 and the error sign value from the error signregister 603 and generates a set of update values, UD(0)–UD(4) thatindicate the sign of the product of the error sign value and the datasign value. The update values are provided to the power scaling logic611 which operates similarly to the embodiments described above inreference to FIGS. 7–9 to generate an updated, power-scaled set ofpre-emphasis tap weights 226.

As discussed in reference to FIG. 19, the threshold values for thepositive and negative LSB sampler (i.e., T+ and T−) may be set tonormalized +/−⅔ signal levels, respectively, which are binary multiplesof the normalized +⅓ threshold level. Accordingly, in the embodiment ofFIG. 22, the adaptive module 600 generates an adaptive threshold controlvalue, C_(TA), that corresponds to the normalized +⅓ signal level, andgenerates control values, C_(T+) and C_(T−) for the positive andnegative LSB samplers, by multiplying C_(TA) by 2 and −2, respectively(i.e., C_(T+)=2C_(TA) and C_(T−)=−2C_(TA)). More specifically, thefinite state machine 607 receives the most recently stored sample value(i.e., MSB_(n−1)/LSB_(n−1)) from the MSB and LSB registers 601 and 603,and asserts an update threshold signal 612 (UT) if the sample valuecorresponds to the +⅓ signal level (i.e., sample value=‘11’). The updatethreshold signal 612 is provided to a count enable input (i.e., strobeinput) of the threshold counter 613, and the error sign value stored inregister 603 is coupled to the up/down input of the threshold counter613. By this arrangement, when the update threshold signal 612 isasserted (indicating that the sample value is a ‘11’), the thresholdcontrol value, C_(TA), maintained within threshold counter isincremented in response to a positive error sign value (i.e., thepositive error sign value indicating that the input signal that yieldedthe n−1 sample value is above the +⅓ level) and decremented in responseto a negative error sign value. In one embodiment, the threshold controlvalue, C_(TA), is supplied to a current DAC within an adaptive sampleras described above in reference to FIGS. 10–12. Alternatively, a DAC maybe provided within the adaptive module 600 to generate an analogthreshold, TA. The multiplier circuit 617 multiplies C_(TA) by 2 (e.g.,by actively or passively shifting the C_(TA) value left by one bit) togenerate a control value for the T+ threshold, C_(T+). The thresholdinverter 619 is provided to flip the sign of C_(T+) to generate C_(T−),the control value for the T− threshold. Thus, the adaptive module 600may be used to adaptively generate the control values applied toestablish sampling thresholds within the positive and negative LSBsamplers and the adaptive sampler of a multi-level, multi-samplereceiver.

As in the embodiment of FIG. 7, the finite state machine 607 asserts anupdate weight signal 610 (UW) to prompt the power scaling logic 611 togenerate an updated set of tap weights 226. In an embodiment in whichthe error sign value corresponds to a logic ‘11’ sample value, thefinite state machine 607 asserts the update weight signal after the MSBregister 601 has been fully loaded (or re-loaded) and the most recentlystored sample value (MSB_(n−1)/LSB_(n−1)) is a logic ‘11’.

In the embodiment of FIG. 22, one or more bits of a multi-bit controlsignal 608 are asserted to initiate an offset calibration operationwithin the adaptive module 600, with the bit (or combination of bits)indicating the sampler to be calibrated (e.g., positive or negative LSBsamplers, MSB sampler or adaptive sampler). The error signal multiplexer621 has a control port coupled to receive a select signal (SEL) from thefinite state machine 607, and four input ports coupled to receiveMSB_(n−1), /LSB_(n−1) (the complement LSB value generated by inverter622), LSB_(n−1) and error sign value, sgn(e_(n−1)), respectively. If theMSB sampler is to be calibrated, a null signal is generated at the MSBsampler input (e.g., by configuring the transmit circuit to transmit anull differential signal, or by switchably coupling the inputs of theMSB sampler to one another) and the most recently stored MSB is selectedby the error signal multiplexer (i.e., in response to the select signal,SEL, from the finite state machine 607) as the offset error signal 624supplied to the up/down input of the offset counter 615. (Also, as shownin FIG. 22, a filter 625 may optionally be provided to filter transientstates in the offset error signal 624). By this operation, if the MSBsampler generates a stream of positive sample values (e.g., MSB=1) inresponse to the null signal input, then the MSB sampler has a negativeDC offset which may be canceled by a positive offset cancellation value.In one embodiment, the finite state machine 607 asserts an update offsetsignal 614 after a predetermined number of samples have been received(e.g., enough samples to establish a stable, filtered signal at theup/down input of the offset counter 615), thereby incrementing theoffset count within the offset counter 615 if the filtered MSB (i.e.,output of filter 625) is positive, and decrementing the offset count ifthe filtered MSB is negative. The output of the offset counter 615 maybe provided to the finite state machine 607, as described in referenceto FIG. 7, to enable detection of a dithering condition within theoffset counter 615 (i.e., indicating convergence to the desired MSBoffset count).

In the embodiment of FIG. 22, the update offset signal 614 is suppliedto the input of the demultiplexer 623 which, in turn, passes the updateoffset signal 614 to the load-enable input of a selected one of offsetregisters 627, 629, 631 and 633 according to the state of the selectsignal, SEL, generated by the finite state machine 607. Parallel loadports of the offset registers 627, 629, 631 and 633 are coupled toreceive the offset count 628 output from the offset counter 615. Thus,during an offset calibration operation on the MSB sampler, eachassertion of the update enable signal 614 results in the offset count628 being loaded (i.e., strobed) into the MSB offset register 627. Bythis operation, when the offset count begins to dither, the updateoffset signal 614 may be asserted a final time to load the desired MSBoffset count into the MSB offset register. In one embodiment, the MSBoffset register is coupled to provide the MSB offset value to a currentDAC within the MSB sampler (e.g., as shown in FIGS. 10 and 12.Alternatively, the MSB offset value may be converted to an analog signalthat is provided to the MSB sampler.

In one embodiment, offset cancellation operations are performed for theremaining samplers (i.e., the positive and negative LSB samplers and theadaptive sampler) in generally the same manner as the MSB sampler,except that the threshold control values provided to the sampler beingcalibrated are temporarily zeroed to enable detection of the DC offset,if any, then restored when the offset calibration operation is complete.Also, in the case of the positive LSB sampler, a logic ‘1’ LSB indicatesa negative L+ sample, and a logic ‘0’ LSB indicates a positive L+sample; a correlation that is the complement of the MSB case (i.e., inwhich a logic ‘1’ MSB corresponds to a positive MSB sample). Inverter622 is provided to account for this complement condition, causing theoffset counter 615 to be incremented in response to a logic ‘0’ L+sample during calibration of the positive LSB sampler.

Clock Recovery

FIG. 23 illustrates an embodiment of a multi-sample, 4-PAM receiver 640that recovers both data and clocking information from the incomingmulti-level signal, x′_(n). The receiver 640 includes a multi-levelsampler 541 (a 4-PAM sampler in this example), adaptive sampler 543,edge sampler 641, adaptive module 643 and clock recovery circuit 645.The 4-PAM sampler 541, adaptive sampler 543 and adaptive module 643operate generally as described in reference to FIG. 19 to generate datasamples 642 (i.e., MSB and LSB) and error samples 218, and to adaptivelyupdate the transmit pre-emphasis tap weights, (226) and the samplerthresholds 550, 552 and 554 (TA, T+ and T−, respectively).

The clock recovery circuit 645 generates a sampling clock signal 210(SCLK) and edge clock signal 610 (ECLK) in response to transitionsamples 644 (T_(n−1)), generated by the edge sampler 641, and the datasamples 642 generated by the 4-PAM sampler 541. In one embodiment, thesampling clock signal 210 is provided to the 4-PAM sampler 541 andadaptive sampler 543 to control the sampling instant therein (as shown,for example, in FIGS. 10 and 12) and thereby define each successive datareception interval. In one embodiment, transitions in the sampling clocksignal 210 are phase aligned with midpoints in the incoming data eyes(i.e., midpoint of data valid intervals in the incoming data signal,x′_(n)), for example, as shown in FIG. 4. In an alternative embodiment,the sampling clock signal 210 may be offset from the midpoints in theincoming data eyes, for example, to accommodate asymmetric setup andhold time requirements in the 4-PAM sampler 541 and/or adaptive sampler543. While only a single sampling clock signal 210 is shown in FIG. 23,multiple sampling clock signals may be generated by the clock recoverycircuit 645 to enable receipt of multi-data rate signals. For example,in a double data rate system, the clock recovery circuit 605 maygenerate SCLK and /SCLK to enable capture of data and error samples inboth odd and even phases of the sampling clock signal 210.

The clock recovery circuit 605 adjusts the phase of the edge clocksignal 610 to maintain phase alignment between the edge clock signal 610and transition points between incoming data eyes. That is, the edgeclock signal 610 is adjusted for edge alignment with data validintervals in the incoming data signal, x′_(n). The edge clock signal 610is supplied to the edge sampler 641 where it is used to time thesampling of transitions in the incoming data signal. One or more storagecircuits (not specifically shown in FIG. 23) may be provided within theedge sampler 641 to latency-align the transition sample, T_(n−1), withthe data sample, MSB/LSB_(n−1) so that, for each pair of successive datasamples 642 supplied to the clock recovery circuit 645 by the 4-PAMsampler 541, the edge sampler 641 supplies a transition sample 644 thatcorresponds to the intervening transition in the incoming signal,x′_(n), if any.

FIG. 24 illustrates possible signal transitions between successive 4-PAMdata transmissions 660 and 662. As shown, from each of four possiblesignal levels, the incoming data signal may transition to any of threeother signal levels. For example, a signal level above T+ (correspondingto data value ‘10’) may transition to (1) a signal level between the T+and zero thresholds (‘10’→‘11’); (2) a signal level between the zero andT− thresholds (‘10’→‘01’); and a signal level below T− (‘10’→‘00’).Examining the different possible transitions, it can be seen that anytransitions that cross all three threshold levels will cross the zerothreshold level at the timing center, T1, between the desired datasampling instants; the desired edge clock transition time. Similarly,transitions that cross a single threshold level will cross either thezero threshold level, the T+ threshold level or the T− threshold levelat T1. By contrast, any transitions that cross two threshold levels, butnot three, do not cross the zero, T+ or T− threshold levels at T1.Enumerating the different transitions that cross the zero, T+ and T−threshold levels at T1 as transition types (1), (2) and (3),respectively, it can be seen that type-1 transitions are those in whichthe LSB remains unchanged at either ‘1’ or ‘0’, while the MSB changesstate (i.e., (MSB_(N)×or MSB_(N−1)) & (LSB_(N)× nor LSB_(N−1))); type-2transitions are those in which the MSB remains high while the LSBchanges state (i.e., MSB_(N) & MSB_(N−1) & (LSB_(N)× or LSB_(N−1))); andtype-3 transitions are those in which the MSB remains low, while the LSBchanges state (i.e., /MSB_(N) & /MSB_(N−1) & (LSB_(N)× or LSB_(N−1))).

In the embodiment of FIG. 23, the clock recovery circuit 645 evaluatessuccessive MSB/LSB values to determine when a type-1 signal transitionhas occurred, and adjusts the phase of the edge clock signal 610 andsampling clock signal 210 according to the state of the correspondingtransition sample 644. In the case of a rising edge transition in theincoming signal, x′_(n) (i.e., ‘00’→‘10’, or ‘0’→‘11’), a logic ‘1’transition sample 644 indicates that the edge clock transition occurredafter the incoming signal transition (i.e., edge clock lags the signaltransition) and therefore that the phase of the edge clock signal 610 isto be advanced. Conversely, a logic ‘0’ transition sample 644 indicatesthat the edge clock transition occurred prior to the incoming signaltransition (i.e., edge clock leads the signal transition) and thereforethat the phase of the edge clock signal 610 should be delayed. The clockrecovery circuit 605 receives the transition samples 644 from edgesampler 641 and data samples from the 4-PAM sampler 642 and adjusts thephase of the edge clock signal 610 as necessary to maintain alignmentbetween the edge clock signal 610 and transitions in the incomingsignal, x′_(n). In one embodiment, the sampling clock signal 210 ismaintained at a substantially constant phase offset from the edge clocksignal 610 such that phase alignment between the edge clock signal 610and data signal transitions yields a desired phase alignment between thesampling clock signal 210 and midpoints in the incoming data eyes.

FIG. 25 illustrates an embodiment of a clock recovery circuit 670 thatadjusts the phase of edge clock signal 610 and sampling clock signal 210based on selected transitions detected in the incoming signal, x′_(n),and that may be used to implement the clock recovery circuit 645 of FIG.23. The clock recovery circuit 670 includes a transition logic circuit671, early/late counter 683, majority detector 685, interpolator 687 andreference loop 689. In the embodiment of FIG. 25, the transition logic671 asserts a transition detect signal 672 (TDET) upon detecting atype-1 transition in a successive pair of data samples, MSB/LSB_(n−2)and MSB/LSB_(n−1), and asserts an early/late signal 674 according to thedirection of the incoming signal transition (rising or falling edge) andthe state of the corresponding transition sample, T_(n−1). Thetransition detect signal 672 is applied to a count enable input (CEN) ofthe early/late counter 683 to enable an early/late count value to beincremented or decremented according to the state of the early/latesignal 674. In one embodiment, the transition logic 671 outputs a logichigh early/late signal 674 if the transition sample, T_(n−1), does notmatch the MSB of the trailing data sample, MSB_(n−1), and a logic lowearly/late signal 674 if the transition sample matches the MSB of thetrailing data sample. That is, if the transition sample, T_(n−1), iscaptured after the transition from MSB/LSB_(n−2) to MSB/LSB_(n−1), thetransition sample will match the MSB_(n−1) sample and thereby indicatethat the edge clock signal transition is late relative to the incomingsignal transition. Conversely, if the transition sample is capturedbefore the transition from MSB/LSB_(n−2) to MSB/LSB_(n−1), thetransition sample will not match the MSB_(n−1) sample, therebyindicating that the edge clock signal transition is early relative tothe incoming signal transition. The following table illustratesexemplary combinations of incoming signal samples (and correspondingtransition type) and transition samples; the resulting transition detectand early/late signals generated by the transition logic circuit 671;and the resulting adjustments to the early/late count and phase of theedge clock sampling clock signals:

TABLE 5 Trans. Early E/L Cnt ECLK/SCLK MSB/LSB_(n−2) MSB/LSB_(n−1)T_(n−1) Type TDET (/Late) Adj. Phase Adjust 00 10 0 1 1 1 +1 Delay 00 101 1 0 0 −1 Advance 01 11 0 1 1 1 +1 Delay 01 11 1 1 0 0 −1 Advance 11 010 1 0 0 −1 Advance 11 01 1 1 1 1 +1 Delay 10 00 0 1 1 0 −1 Advance 10 001 1 1 1 +1 Delay 11 10 X 2 0 X 0 No change 10 11 X 2 0 X 0 No change 0001 X 3 0 X 0 No change 01 00 X 3 0 X 0 No change 00 11 X — 0 X 0 Nochange 01 10 X — 0 X 0 No change 11 00 X — 0 X 0 No change 10 01 X — 0 X0 No change

In one embodiment, the early/late counter 683 is initialized to zeroand, as illustrated in Table 5, is incremented in response to an earlyindication (i.e., a logic high early/late signal 674) and decremented inresponse to a late indication (i.e., a logic low early/late signal 674).By this operation, the sign bit (e.g., the MSB) of the early/late countmaintained within the early/late counter 683 indicates whether moreearly than late indications, or more late than early indications havebeen received from the transition logic 671 (i.e., the count value willunderflow to a negative value if more late indications than earlyindications are detected). Accordingly, after a predetermined number oftransition detect assertions (or after a predetermined time), themajority detector 685 evaluates the sign of the early/late count (i.e.,signal 684) and outputs an up/down signal 688 to the interpolator 687accordingly. The early/late count value may then be reset to zero inpreparation for counting a subsequent set of early/late indications.

In one embodiment, the interpolator 687 maintains an interpolationcontrol word that is incremented in response to a logic high up/downsignal 688 and decremented in response to a logic low up/down signal688. The most significant bits of the interpolation control word areused to select a pair of phase vectors from the set of N phase vectors692 generated by the reference loop 689, and the least significant bitsof the interpolation control word are used to interpolate between theselected pair of phase vectors. As the control word is incremented, theinterpolation is incrementally shifted from a leading one of the phasevectors to a lagging one of the phase vectors, thereby incrementallydelaying (i.e., retarding) the phase of the edge and sampling clocksignals 610, 210. Conversely, as the control word is decremented, theinterpolation is incrementally shifted toward the leading one of theselected phase vectors, thereby incrementally advancing the phase of theedge and sampling clock signals 610, 210.

In one embodiment, the reference loop 689 is formed by a delay lockedloop (DLL) that receives a reference clock signal 690 and, in response,generates a plurality of phase vectors 692 that are phase distributedwithin a cycle time of the reference clock signal 690. Alternatively,the reference loop 689 may be a phase locked loop (PLL) that multipliesthe reference clock frequency to generate a plurality of phase vectors692 having a higher frequency than the reference clock frequency. Inanother alternative embodiment, the reference loop 689 may include aninternal timing reference generator (e.g., a ring oscillator or otherclock generating circuit) so that no reference clock signal 690 isrequired. Also, as discussed above, the interpolator 687 may generateany number of sampling clock and edge clock signals. For example, in adouble data rate system, the interpolator 687 generates an edge clocksignal and complement edge clock signal, and a sampling clock signal andcomplement sampling clock signal, the sampling clock signal being offsetfrom the edge clock signal by a quarter cycle (90 degrees) of the edgeclock signal. The quarter cycle offset may be achieved, for example, bya second interpolator that maintains a control word having a 90 degreedigital offset from the control word used to generate the edge clocksignal. Other techniques may be used to generate the edgeclock-to-sampling clock offset in alternative embodiments. In a quaddata rate system, the interpolator 687 (or multiple interpolators)generates four edge clock signals and four sampling clock signals, thecombined set of eight clock signals being evenly offset in phase over acycle time of the edge clock signal (i.e., 45 degree increments betweensuccessive clock edges). This approach may be extended to supportvirtually any data rate.

It should be noted that numerous changes may be made to the clockrecovery circuit 670 of FIG. 25 without departing from the scope of thepresent invention. For example, in one alternative embodiment, theup/down signal 688 is a two-bit signal in which the ‘00’ state signals ahold condition. The interpolator 687 responds to the hold condition bymaintaining the interpolation control word at its present value. In suchan embodiment, the majority detector 685 may receive the entireearly/late count from the early/late counter, and output the up/downsignal in the ‘00’ state if the count value indicates a balancedreception of early and late detections (e.g., the early/late count iszero). Alternatively, the majority detector 685 may be omittedaltogether and the sign of the early/late count value output directly tothe interpolator 687 to control the phase adjustment of the edge andsampling clock signals 610 and 210.

Returning to FIG. 24, it can be seen that the type-2 and type-3transitions cross the T+ and T− thresholds, respectfully, in synchronismwith the desired transition time of the edge clock signal 610 (i.e.,T1). Consequently, the type-2 and type-3 transitions may be detected andused along with, or instead of, the type-1 transitions to recover theedge and sampling clock signals 610 and 210. In one embodiment,additional edge samplers 641 are provided to generate transition samplesat the T+ and/or T− thresholds. Additional circuitry is also providedwithin the clock recovery circuit 670 of FIG. 25 to detect the 11-to-01and/or 00-to-10 transitions and, in response, to update the early/latecounter 683 according to the corresponding transition samples. By thisarrangement, the overall number of incoming signal transitions used forclock recovery is increased, thereby relaxing the transition densityrequired in the incoming signal for clock recovery purposes.

Returning to FIG. 23, threshold multiplexers and output pathmultiplexers similar to multiplexers 507, 509, 511 and 513 of FIG. 18may be provided to enable the adaptive sampler 543 to proxy for any ofthe component samplers of the 4-PAM sampler 541. By this operation,component samplers of the 4-PAM sampler 541 may be taken out of serviceone at a time and calibrated (e.g., offset cancellation calibration),tested or used for other purposes. Also, if the adaptive sampler 543exhibits improved performance relative to one of the component samplersof the 4-PAM receiver, the adaptive sampler 543 may be substituted forthe component sampler during normal operation.

In the embodiment of FIG. 23, the adaptive sampler 543 is clocked by thesampling clock signal 210 and therefore captures samples at the sametime as the component samplers of the 4-PAM sampler 541. In analternative embodiment, the adaptive sampler 543 may be clocked by aselectable-phase clock signal having an independently selectable phaseoffset. By alternately switching the phase of the selectable-phase clocksignal to match the phase of the sampling clock signal 210 and the edgeclock signal 610, the adaptive sampler 543 may be used as a proxysampler for the component samplers of the 4-PAM sampler 541 as well asthe edge sampler 641. Also, if one of the edge samplers may be taken outof service (e.g., in a mesochronous or plesiochronous system having afrequency offset estimation (via a second order feedback loop, forexample), the edge sampler may be used as a proxy for an adaptivesampler (if provided), data sampler or other sampler within thereceiver. Further, while a 4-PAM system is described in reference toFIG. 23, edge samplers may be used for clock recovery purposes in binarysignaling systems (or multi-level signaling systems having more thanfour signal amplitude levels). In such systems, the edge samplers may beused as proxy samplers for adaptive and/or data samplers.

Transmit equalization can cause multi-modal distributions in edgecrossings. This in turn causes the conventional clock-data-recovery loopto produce less accurate estimates on the phase of the incoming datastream. In one embodiment, error signals at both data and edge samplesare combined to form the update of the equalizer taps, thereby reducingloss of timing accuracy in effect by trading off between timing accuracyand voltage accuracy due to equalizer compensation. The use of data andedge error signals to update equalizer taps are illustrated, for exampleand without limitation, by the update expression:

-   W _(N+1)=W _(N+)stepsize_(wd)*sign(e_(dn))*sign(u    _(dn))+step_(we)*sign(e_(en))*sign(u_(en)), where stepsize_(wd) is a    data-weighted update factor and stepsize_(we) is an edge-weighted    update factor. The subscript “dn” refers to the n^(th) data sample    and the subscript “en” refers to the n^(th) edge sample. As    discussed above in reference to FIG. 16, in a single phase tap    weight update operation, x′_(n) may be used in place of u_(dn) and    edge samples edge_(n) (e.g., obtained by filter for edge transitions    such as when x_(n)+x_(n−1)=0) may be used in place of u_(en).    Alternatively, if one of the edge samplers may be taken out of    service (e.g., in a mesochronous or plesiochronous system having a    frequency offset estimation (e.g., via a second order feedback    loop), the edge sampler may be used as a proxy for an adaptive    sampler (if provided), data sampler or other sampler within the    receiver.

The term including the error from the data samples guides the equalizerupdates toward the negative gradient direction of the mean-square-erroron data samples, while the term including the error from edge samplesguides the equalizer updates toward the negative gradient direction ofthe mean-square-error on edge samples. Said differently, the termincluding error in data samples affects the equalizer such that it makesthat error smaller, while the term including error in edge samplesaffects the equalizer such that it makes the error at the edges smaller.In case when there are competing effects between these two errors, theequalizer is able to achieve the balance. This tradeoff may be achievedwith different relative magnitude of step sizes (weighting) for data andedge errors.

A convenient aspect of the embodiments of FIGS. 23 (and 26 describedbelow) is that the clock recovery loop already generates the edge errorsignals and conveniently filters them (i.e. generates them) only onvalid transitions (i.e., by detecting early-late signals as discussedabove). Hence, little or no additional circuitry in the receiver isneeded to generate the edge error signals.

FIG. 26 illustrates a double-data-rate, multi-sample receiver 700according to an embodiment of the invention. The receiver 700 includes4-PAM samplers 701 ₁–701 ₄, data/edge sample deserializer 704, adaptivesamplers, 703 ₁ and 703 ₂, error sample deserializer 709, an adaptivemodule 705 and clock recovery circuit 707. Each of the 4-PAM samplers701 ₁–701 ₄ operates in generally the same manner as the multi-levelsampler 541 of FIG. 19, and includes an MSB sampler 563 to compare anincoming signal, x′_(n), with a zero threshold, and positive andnegative LSB samplers 561 and 565 to compare the incoming signal withadaptively generated thresholds, T+ and T− (e.g., adapted to thenormalized +⅔ signal levels). Two of the 4-PAM samplers 701 ₁ and 701 ₃are used to generate two-bit data samples (i.e., each sample having andMSB and LSB) in response to odd and even sampling clock signals, CLK_DOand CLK_DE, respectively. The remaining two 4-PAM samplers, 701 ₂ and701 ₄, are used to generate transition samples in response to odd andeven edge clock signals (CLK_EO and CLK_EE), with the MSB sampler beingused to detect type-1 data signal transitions, and the positive andnegative LSB samplers being used to detect type-2 and type-3 data signaltransitions, respectively. The data and edge sample values generated bythe 4-PAM samplers 701 ₁–701 ₄ are supplied to the data/edge sampledeserializer 704, which shifts the incoming serial stream of MSB and LSBsamples (after performing LSB+× or LSB−) and transition samples intorespective shift registers. The contents of the shift registers withinthe data/edge deserializer 704 constitute parallel words of MSBs, LSBsand transition samples (i.e., MSB[N:0], LSB[N:0] and T[M:0],respectively, where M≦N due to the fact that not all transitions aretype-1, type-2 or type-3 transitions) that are supplied to the clockrecovery circuit 707 and adaptive module 705. The clock recovery circuit707 operates generally in the manner described in reference to FIGS.23–25 to generate even and odd edge and data clock signals, CLK_EE,CLK_EO, CLK_DE and CLK_DO (e.g., the even and odd clock signals beingcomplements of one another, and the edge and data clock signals beingquadrature-offset from one another). The adaptive module 705 applies theincoming data samples in tap weight update operations to generatepower-scaled, updated tap weights W_(N+1) and, when instructed, toperform offset cancellation operations as described in reference to FIG.22 for the component samplers within each of the 4-PAM samplers 701₁–701 ₄. For example, the adaptive module 705 generates three offsetcancellation values, OFST_DO(3), for the odd-data 4-PAM sampler 701 ₁ inthe manner described in reference to FIG. 22, and similarly generatesoffset cancellation values OFST_DE(3), OFST_EO and OFST_EE, for theeven-data 4-PAM sampler 701 ₃, odd-edge 4-PAM sampler 701 ₂ andeven-edge 4-PAM sampler 701 ₄.

In the embodiment of FIG. 26, the adaptive samplers 703 ₁ and 703 ₂ areclocked by respective odd and even adaptive-sampler clock signals,CLK_AO and CLK_AE, and generate error samples by comparing the incomingsignal, x′_(n), with adaptive sampler thresholds T_AO and T_AE,respectively. In one embodiment, the adaptive module 705 iterativelyadjusts each of the adaptive sampler thresholds (i.e., in response tothe incoming error samples, ERR[N:0], or a subset thereof) to thenormalized +⅓ signal level and uses the adaptive-sampler threshold asdiscussed above in reference to FIG. 22 to generate the T+ and T−thresholds supplied to the 4-PAM samplers (e.g., doubling the adaptivesampler threshold to generate T+, then complementing T+ to generate T−).The error samples generated by the adaptive samplers 703 ₁ and 703 ₂ areprovided to the error sample deserializer 709 which shifts the odd- andeven-phase error samples (i.e., the error samples alternately generatedby adaptive samplers 703 ₁ and 703 ₂) into a shift register for paralleldelivery to the adaptive module (i.e., ERR[N:0]).

In one embodiment, the odd and even adaptive-sampler clock signals aregenerated by respective interpolators within the clock recovery circuit707, and therefore have independently selectable phase offsets. By thisarrangement, clock signal CLK_AO may be selectively phase aligned witheither of the odd-phase data and edge clock signals, CLK_DO and CLK_EO,so that adaptive sampler 703 ₁ may proxy for any of the componentsamplers within the odd-phase 4-PAM data sampler 703 ₁, and any of thecomponent samplers within the odd-phase 4-PAM edge sampler 703 ₂.Similarly, clock signal CLK_AE may be selectively phase aligned witheither of the even-phase data and edge clock signals, CLK_DE and CLK_EE,so that adaptive sampler 703 ₂ may proxy for any of the componentsamplers within the even-phase 4-PAM data sampler 703 ₃, and any of thecomponent samplers within the even-phase 4-PAM edge sampler 703 ₄. Inalternative embodiments, each of the adaptive samplers may proxy for anycomponent sampler within any of the 4-PAM samplers. By this arrangement,one of the adaptive samplers 703 may continue to generate the errorsamples needed to adaptively update the pre-emphasis tap weights, W_(N+1), and the thresholds T_AO and T_AE (and, by extension, the T+ andT− thresholds), while the other of the adaptive samplers 703 is used asa proxy sampler for a component sampler of one of the 4-PAM samplers701. The adaptive module 705 additionally generates an offsetcancellation value for each of the adaptive samplers 703 (i.e., OFST_AOand OFST_AE), for example, by nulling the input to the adaptive sampler,zeroing the threshold of the adaptive sampler, and adjusting the offsetcancellation value for the adaptive sampler until the error samplesgenerated by the adaptive sampler begin to dither.

FIG. 27 illustrates a portion of the receiver 700 of FIG. 26 in greaterdetail, showing the threshold multiplexers and output path multiplexersthat may be used to enable the odd-phase adaptive sampler 703 ₁ to be aproxy sampler for any of the component samplers 561, 563 and/or 565within the 4-PAM data sampler 701 ₁ or 4-PAM edge sampler 701 ₂. Asimilar set of threshold multiplexers and output path multiplexers maybe coupled to the even-phase adaptive sampler 703 ₂ and 4-PAM data andedge samplers 701 ₃ and 701 ₄.

Referring to 4-PAM sampler 701 ₁, threshold multiplexer 725 is providedto select either the T+ threshold or the adaptive sampler threshold,T_AO, to be summed with the offset cancellation OFSC_DO[2] and providedto the positive LSB sampler 561. Similarly, threshold multiplexer 729 isprovided to select either the T− threshold or the adaptive samplerthreshold, T_AO, to be summed with offset cancellation OFSC_DO[0] andprovided to the negative LSB sampler 565, and threshold multiplexer 727is provided to select either the zero threshold or the adaptive samplerthreshold T_AO, to be summed with offset cancellation OFSC_DO[1] andprovided to the MSB sampler 563. Output multiplexers 735, 737 and 739are provided in the 4-PAM sampler 701 ₁ to select either the output ofthe odd-phase adaptive sampler 703 ₁ or the output of the componentsamplers 561, 563 and 565, respectively, to be provided to the data/edgesample deserializer 704. Threshold multiplexers 725, 727 and 729, andoutput multiplexers 735, 737 and 739 are provided within the odd-phaseedge sampler 7012 and coupled to the component samplers thereof in thesame way that like-numbered multiplexers are coupled to the componentsamplers of the odd-phase data sampler 701 ₁.

Threshold multiplexer 730 is provided to enable any of the T+, 0, T− andT_AO threshold levels to be summed with offset cancellation OFSC_AO andprovided to the adaptive sampler 543 (i.e., sampler 543 being thesampling circuit within the overall sampler 703 ₁). Output pathmultiplexer 731 is provided to select the output of any one of thecomponent samplers of 4-PAM samplers 701 ₁ and 701 ₂ or the adaptivesampler 543 to be provided to the error sample deserializer 709. By thisarrangement, the adaptive sampler 543 may operate as a proxy sampler forany of the component samplers of the odd-phase data and edge samplers701 ₁ and 701 ₂, and vice-versa, thereby enabling calibration operationsor other out-of-service operations to be performed on the odd-phase dataand edge samplers without interrupting the recovered stream of data andedge samples. As discussed in reference to FIG. 18, the threshold andoutput path multiplexers may be independently controlled to enable amake-before-break transition between a component sampler (i.e., 561, 563or 565) and the adaptive sampler 543, establishing the alternate sourceof sample values before taking the component sampler out of service. Theeven-phase data, edge and adaptive samplers (i.e., 701 ₃, 701 ₄ and 703₂, respectively) may include threshold multiplexers and output pathmultiplexers coupled in the same manner as the threshold multiplexersand output path multiplexers shown for odd samplers in FIG. 27.

Still referring to FIG. 27, the odd-phase adaptive sampler 703 ₁receives the phase-selectable clock signal, CLK_AO, and therefore maygenerate sample values in phase with either the odd-phase data clocksignal, CLK_DO, or the odd-phase edge clock signal, CLK_EO. Theeven-phase adaptive sampler similarly receives the phase-selectableclock signal, CLK_AE, and therefore may generate sample values in phasewith either the even-phase data clock signal, CLK_EO, or the even-phaseedge clock signal, CLK_EE.

Dual Mode, Multi-PAM Receiver

In one embodiment, the 4-PAM sampler illustrated in FIG. 19 may beselectively operated in either a 2-PAM mode (i.e., binary signaling) ora 4-PAM mode, according to application needs and/or signaling systemcharacteristics. For example, the 2-PAM mode may be selected upondetermining that signaling margins in a given system are insufficientfor 4-PAM signal resolution. Also, a signaling system may be dynamicallyswitched between 4-PAM and 2-PAM modes as signaling characteristicsdictate, or to allow one or more of the component samplers of the 4-PAMsampler to be taken out of service (e.g., for calibration purposes) orto allocate one or more of the component samplers to a differentfunction.

FIG. 28 illustrates an embodiment of a multi-sample, multi-levelreceiver 765 in which the positive and negative LSB samplers 561 and 565of a 4-PAM sampler 541 are used as adaptive samplers when the 4-PAMsampler 541 is operated in a 2-PAM mode. As in the embodiment of FIG.19, the incoming signal, x′_(n) is supplied to all three componentsamplers of the 4-PAM sampler 541. The positive LSB sampler 561 comparesthe incoming signal with the T+ threshold and generates a correspondingerror sign value, sgn(e_(H)), that indicates whether the incoming 2-PAMsignal exceeds the T+ threshold. The negative LSB sampler 565 similarlycompares the incoming signal with the T− threshold and generates acorresponding error sign value, sgn(e_(L)), that indicates whether theincoming signal exceeds the T− threshold. When a live enable signal 772is in a logic ‘1’ state, a live adaptation mode is selected within thereceiver 765. In the live adaptation mode, pre-emphasis tap weights andreceiver threshold levels are iteratively updated using error signalsgenerated from live rather than predetermined data sequences). Morespecifically, the live enable signal 772 is provided to a control inputof multiplexer 773 so that, when the live adaptation mode is selected,the multiplexer 773 outputs the MSB sample generated by MSB sampler 563(i.e., the sign of the incoming 2-PAM signal) to the control input ofmultiplexer 770. Multiplexer 770, in response, selects either thepositive or negative LSB sampler (i.e., 561 or 565) to provide an errorsample 774 to an adaptive module 771. Thus, when the incoming 2-PAMsignal is positive, the error sign value generated by the positive LSBsampler 561 is selected for use in a tap weight update operation (and T+threshold update), and when the incoming 2-PAM signal is negative, theerror sign value generated by the negative LSB sampler 565 is selectedfor use in a tap weight update operation (and T− threshold update).Thus, the sign of the 2-PAM sample value generated by the MSB sampler563 is used to select the appropriate error source in each receptioninterval, thereby enabling the T+ and T− thresholds to be adapted to thecorresponding high and low levels of the 2-PAM signal, and enabling morerapid gathering of error information for use in tap weight updates.

When the live enable signal 772 is deasserted, a batch update mode isselected, and the sign of the originally transmitted data value, x_(n),is used to select either the positive LSB sampler 561 or negative LSBsampler 565 to provide the error sample 774 to the adaptive module. Asdiscussed above, in batch mode, the sign of the transmitted data valuemay be known at the receive-side IC device, for example, by sending thedata transmission sequence in advance of the batch update operation, orby storing the transmit data pattern in both the transmit- andreceive-side devices. In either case, the error sign values generated bythe positive and negative LSB samplers 561 and 565 may be applied in thesame manner as in the live adaptation mode to adapt the T+ and T−thresholds to the upper and lower binary signal levels, and to updatethe pre-emphasis tap weights.

Alternative Indicator Functions

In the signaling system embodiments described above, error samplesgenerated by an adaptive sampler within a multi-sample receiver areapplied to update transmit pre-emphasis tap weights in repeatedsign-sign LMS update operations. Because the adaptive sampler generateserrors with respect to an expected data level, logical filtering of datais used to ensure that the incoming signal in fact corresponds to theexpected data level. For example, in a binary signaling embodiment inwhich the adaptive sampler receives a target data level threshold thatcorresponds to a logic ‘1’ data transmission, the error sample generatedby the adaptive sampler is applied in a tap weight update if thecorresponding data sample is a logic ‘1’. Similarly, in a multi-PAMsignaling embodiment, the error sample is applied in a tap weight updateoperation if the corresponding data sample corresponds to the adaptivesampler threshold level (e.g., +⅓ the normalized signal level in theembodiment of FIG. 19). In effect, the logical filtering of incomingdata samples constitutes an indicator function that may be expressed aspart of the sign-sign LMS operation. For example, indicator functionsfor the 2-PAM (i.e., binary) and 4-PAM signaling systems described inreference to FIGS. 3 and 19 may be expressed as follows:I _(LMS)=(x′ _(n)≧0), (2-PAM; DLEV adapted to logic ‘1’ signal level);I _(LMS)=(T+>x′ _(n)≦0) (4-PAM; TA adapted to logic ‘11’ signal level).These indicator functions may be combined with the update expression (3)above, as follows:W _(N+1) =W _(N) +I _(LMS)·(stepsize*sign(e _(n))*sign( x′))  (9).In alternative embodiments, other indicator functions may be used, andthe indicator function may be omitted altogether, for example, byproviding one or more additional adaptive samplers having thresholds setat all (or a subset) of the expected incoming data levels.

In another alternative embodiment, a trap indicator function is used tofilter errors applied in tap weight update operations (i.e.,update-triggering errors) according to the error magnitude anddirection. Referring to the normalized 2-PAM data eye 801 illustrated inFIG. 29, update-triggering errors are limited to those errors for whichthe corresponding sample value is positive (i.e., sgn(x′_(n))=1), butfalls below the normalized, +1 signal level by more than a thresholdamount. That is, the incoming signal level falls within a trap zonedefined by the zero threshold and a trap threshold, T_(TRP), andtherefore corresponds to a relatively closed data eye. In oneembodiment, illustrated in FIG. 30, the trap threshold, T_(TRP), isadaptively generated by an adaptive module 815 according to the rate oferrors falling within the trap zone, and is supplied to the adaptivesampler 213 as shown in FIG. 30. Overall, the trap indicator functionmay be expressed as follows:(sgn(x′ _(n))=1)&&(sgn(e _(n))=0)   (10),where ‘&&’ denotes a logical AND operation. The error sign value,sgn(en) may be expressed as the sign of the incoming signal less thetrap threshold, so that expression 10 becomes:(sgn(x′ _(n))=1)&&(sgn(x′ _(n) −T _(TRP))=0)   (11), which correspondstoT_(TRP)>x′_(n)≧0   (12).

In one embodiment, the adaptive module 815 adaptively adjusts the trapthreshold to obtain a target count of update-triggering errors per unittime, referred to herein as the target error count. The target errorcount may be a predetermined value that is programmed within thereceive-side IC device (or transmit-side IC device) during run-timeconfiguration or during a production-time programming operation (e.g.,fuse blowing operation, or storage in a nonvolatile memory), orhardwired within the receive-side IC device (or transmit-side ICdevice). In one embodiment, the target error count is initially set to arelatively high number so that the adaptive module 815 drives the trapthreshold higher (thereby increasing the number of incoming signals thatfall within the trap zone) and the trap threshold quickly converges to astable level. After the trap threshold has converged, the target errorcount is lowered (e.g., one time or iteratively) so that fewer errors,having more substantial offset from the normalized +1 signal level, arecounted as errors. The error samples (i.e., sgn(x′_(n)−T_(TRP))) areapplied within the adaptive module 815 along with data sign valuesgenerated by the data sampler 211 in tap weight update operations.

FIGS. 31 and 32 illustrate implementation of a trap zone in a dual mode2-PAM/4-PAM signaling system. As discussed above, when operated in 2-PAMmode, the positive and negative LSB samplers 561 and 565 of a 4-PAMsampler 541 may be idled or used for other purposes. In the embodimentof FIG. 31, the T+ threshold is adjusted to a trap level, T_(TRP+), thatis offset from the normalized +1 signal level, thereby establishing atrap zone between the 0 and adjusted T+ threshold levels. Referringbriefly to FIG. 21, it can be seen that signals falling between the 0and T_(TRP+) thresholds have a logic ‘11’ sample state so the trapindicator function may be expressed as:(MSB=1)&&(LSB=1)   (13).

In one embodiment, illustrated in FIG. 32, the T_(TRP+) threshold isiteratively adjusted by an adaptive module 825 according to the rate oferrors falling within the trap zone, and is supplied to the positive LSBsampler 561. In an embodiment, where the T− threshold is generated bycomplementing the sign of the T+ threshold, the T− threshold becomesT_(TRP−), a threshold offset from the normalized −1 signal level in thesame manner that T_(TRP+) is offset from the normalized +1 signal level.Thus, when the 4-PAM sampler 541 is operated in 2-PAM mode, theotherwise unused positive and negative LSB samplers 561 and 563 may beused to detect signals falling within a trap zone, thereby enabling thetransmit pre-emphasis tap weights to be updated based on errors thatexceed a predetermined, or adaptively generated threshold.

Section headings have been provided in this detailed description forconvenience of reference only, and in no way define, limit, construe ordescribe the scope or extent of such sections. Also, while the inventionhas been described with reference to specific embodiments thereof, itwill be evident that various modifications and changes may be madethereto without departing from the broader spirit and scope of theinvention. Accordingly, the specification and drawings are to beregarded in an illustrative rather than a restrictive sense.

1. A method of operation within a signaling system, the methodcomprising: outputting a first signal from a transmit circuit having aplurality of output drivers; determining whether the first signalexceeds a threshold level; adjusting the threshold level according towhether the first signal exceeds the threshold level; and adjusting adrive strength of at least one output driver of the plurality of outputdrivers according to whether the first signal exceeds the thresholdlevel.
 2. The method of claim 1 wherein outputting a first signal fromthe transmit circuit comprises outputting a plurality of componentsignals from the plurality of output drivers, respectively, theplurality of component signals each contributing to the first signal. 3.The method of claim 2 wherein outputting a plurality of componentsignals from the plurality of output drivers comprises sinking aplurality of currents within the plurality of output drivers to generatea voltage level on a first signal line.
 4. The method of claim 3 whereinsinking a plurality of currents within the plurality of output driversto generate a voltage level on the first signal line comprises drawing atotal current from the first signal line that is a sum of the pluralityof currents, the first signal line being coupled to a reference voltagevia a termination impedance such that the voltage level generated on thefirst signal line is a function of the total current, the terminationimpedance and the reference voltage.
 5. The method of claim 4 whereinthe first signal line has a first end and a second end, and wherein thefirst signal line is coupled to the reference voltage at the first end.6. The method of claim 5 wherein the first signal line is additionallycoupled to the reference voltage at the second end.
 7. The method ofclaim 1 wherein outputting a first signal from a transmit circuitcomprises outputting the first signal onto a signal path, and whereindetermining whether the first signal exceeds a threshold level comprisessampling the first signal within a sampling circuit coupled to thesignal path.
 8. The method of claim 7 wherein sampling the first signalwithin a sampling circuit comprises generating a sample value havingeither a first state or a second state according to whether the firstsignal exceeds the threshold level.
 9. The method of claim 8 whereingenerating a sample value having either a first state or a second statecomprises: comparing the first signal with the threshold level;generating the sample value in the first state if the first signalexceeds the threshold level; and generating the sample value in thesecond state if the threshold level exceeds the first signal.
 10. Themethod of claim 8 wherein adjusting the threshold level comprises:increasing the threshold level if the first signal exceeds the thresholdlevel; and decreasing the threshold level if the threshold level exceedsthe first signal.
 11. The method of claim 7 wherein the first signal isa differential signal having first and second component signals, andwherein outputting the first signal onto the signal path comprisesoutputting the first component signal onto a first signal line of thesignal path and outputting the second component signal onto a secondsignal line of the signal path.
 12. The method of claim 11 whereinsampling the first signal comprises generating a sample value havingeither a first state or a second state according to whether the firstcomponent signal exceeds the second component signal by more than thethreshold level.
 13. The method of claim 12 wherein generating a samplevalue having either a first state or a second state according to whetherthe first component signal exceeds the second component signal by morethan the threshold level comprises biasing a differential amplifierwithin the sampling circuit such that output nodes of the differentialamplifier are driven to substantially the same voltage levels when thefirst component signal exceeds the second component signal by thethreshold level.
 14. The method of claim 12 wherein adjusting thethreshold level comprises: increasing the threshold level if the firstcomponent signal exceeds the second component signal by more than thethreshold level; and decreasing the threshold level if the firstcomponent signal does not exceed the second component signal by morethan the threshold level.
 15. The method of claim 1 wherein adjusting adrive strength of at least one output driver comprises updating aplurality of drive strength values that respectively control drivestrengths of the plurality of output drivers.
 16. The method of claim 15wherein updating a plurality of drive strength values comprisesincrementing a first drive strength value of the plurality of drivestrength values to increase the drive strength of the at least oneoutput driver.
 17. The method of claim 16 wherein updating the pluralityof drive strength values comprises decrementing a second drive strengthvalue of the plurality of drive strength values to decrease the drivestrength of a corresponding one of the plurality of output drivers. 18.The method of claim 16 further comprising adjusting one or more othersof the drive strength values to maintain a sum of the drive strengthvalues within a predetermined maximum value.
 19. The method of claim 18further comprising scaling the drive strength values according to ascaling factor to maintain a sum of the drive strength values within apredetermined maximum value.
 20. A method of operation within anintegrated circuit device, the method comprising: incrementallyadjusting a plurality of drive strength values used to control signallevels generated by a corresponding plurality of output drivers includedwithin a transmit circuit, the adjusted plurality of drive strengthvalues representing a first level of power consumption in the transmitcircuit; determining whether the first level of power consumptionexceeds a maximum level of power consumption in the transmit circuit;and reducing a predetermined one of the plurality of drive strengthvalues if the first level of power consumption exceeds the maximum levelof power consumption.
 21. The method of claim 20 wherein incrementallyadjusting the plurality of drive strength values comprises adjusting asubset of the plurality of drive strength values that excludes thepredetermined one of the plurality of drive strength values.
 22. Themethod of claim 21 wherein the subset of the plurality of drive strengthvalues are provided to a subset of the plurality of output drivers usedto mitigate inter-symbol interference resulting from transmission of asignal by a primary output driver of the plurality of output drivers.23. The method of claim 22 wherein the predetermine one of the pluralityof drive strength values is provided to the primary output driver. 24.The method of claim 20 further comprising: determining whether the firstlevel of power consumption is below a minimum level of power consumptionin the transmit circuit; and increasing the predetermined one of theplurality of drive strength values if the first level of powerconsumption is below the minimum level of power consumption.
 25. Themethod of claim 20 wherein the maximum level of power consumption is aprogrammed value.
 26. The method of claim 20 wherein the maximum levelof power consumption corresponds to a peak power constraint of thetransmit circuit.
 27. The method of claim 20 wherein the maximum levelof power consumption corresponds to an average power constraint of thetransmit circuit.